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> APP 716
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Engineering Topics > APP 716

Aug 21, 2002

Keywords: EMI, EMC, voltage regulator, LDO,
switching regulator, step-down, buck, step-up, boost, flyback,
SEPIC, common-mode, Faraday shield, core geometries, leakage
inductance, crosstalk, ESR, ESL
Related
parts

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APPLICATION NOTE 716

Proper Layout and Component Selection Controls
EMI

Abstract: Understanding
the physics of voltage regulator topologies is important in
designing power systems for EMI and EMC compliance. In
particular, the physical principles behind switching
regulators (buck, boost, flyback, and SEPIC topologies) guide
component choice, magnetics design and PC board layout.
Parasitic elements such as leakage inductance, ESR, and ESL
are significant when optimizing circuit performance.

Most
portable devices include a regulator or other form of power
supply, and the lower supply voltages associated with
smaller-lithography ICs have mandated these power circuits in
many nonportable devices as well. Though not fully understood
by many designers, the trade-off among different types of
regulators and power supplies can have a major effect on
battery life, compliance with electromagnetic
interference/electromagnetic compatibility (EMI/EMC)
regulations, and even the basic operation of a product under
design. The following overview covers the mechanisms and the
physical principles governing the generation and the
propagation of electrical noise in power supplies.

Voltage Regulators
The most common power converter is
the voltage
regulator. It accepts a voltage that varies over a given
range, and it generates an output voltage that does not vary.
Regulators comprise two main categories: switching types and
all others (mainly the linear and shunt types). Unlike
switching regulators, linear and shunt types are limited by
the fact that their output voltage must remain less than the
input voltage. Also, the efficiency of most switching
regulators is better than that of an equivalent linear or
shunt regulator. Nevertheless, the low noise and the
simplicity of linear/shunt types make them an attractive
alternative to switching regulators.

The simplest type
of voltage regulator is a shunt regulator, which merely
adjusts current through a resistor to drop the input voltage
to a regulated output level. Zener diodes also function this
way, but power dissipation in a zener is high, and its load
regulation (change in output voltage with change in load
current) is poor. Some shunt regulators let you set the
regulation voltage with a voltage divider, but those types
usually appear as building blocks in more complex regulators
or power supplies. Generally, shunt regulators are appropriate
for low-power systems in which the variation of load current
is small. This narrow range of application can be expanded,
however, by adding an active pass element (usually a bipolar
transistor) that transforms the shunt into a linear regulator.

Linear Voltage Regulators
Linear voltage regulators
use an active pass element (bipolar or MOSFET)
to drop the input voltage down to the regulated output
voltage. Among these devices, the low-dropout (LDO) types have
become popular over the last decade. Dropout refers to the
minimum difference (between input and output voltage) that
sustains regulation. Dropout voltages as high as 1V have been
called LDO, but a more typical value is between 100mV and
300mV.

Because a linear regulator's input current is
approximately equal to its output current, its efficiency
(output power divided by input power) is a function of the
output/input voltage ratio. Thus, dropout is important, as
lower dropout means higher efficiency. However, if the input
voltage is much higher than the output voltage, or if it
varies widely, then maximum efficiency is difficult to
achieve. Another function of LDO regulators (to be discussed)
is to serve as a barrier to the noise generated by a switching
regulator. In that role, the LDO regulator's low-dropout
characteristic improves the circuit's overall efficiency.

Switching Regulators
If the performance of a linear or
shunt regulator is not adequate for the application, then the
designer must turn to a switching regulator. Along with
improved performance, however, come the drawbacks of larger
size and cost, greater sensitivity to (and generation of)
electrical noise, and a general increase in complexity.

Noise generated by a switching regulator or power
supply can emerge through conduction or radiation. Conducted
emissions can take the form of voltage or current, and each of
these can be further categorized as common-mode or
differential-mode conduction. To complicate matters, the
finite impedance
of connecting wires enables voltage conduction to cause
current conduction and vice versa, and differential-mode
conduction to cause common-mode conduction and vice
versa.

However, generally you can optimize a circuit to
reduce one or more of these emissions. Conducted emission
usually poses a greater problem for fixed systems than for
portable systems. Because the portable device operates from
batteries, its load and source
have no external connections for conducting
emissions.

To understand the source of noise in a
switching regulator, you must first understand its operation.
Descriptions of the many types of switching regulators are
beyond the scope of this article. But, basically, a switching
regulator converts the source voltage/current to load
voltage/current by employing active elements (transistors and
diodes) to shuttle current through storage elements (inductors
and capacitors). To illustrate, the MAX1653 DC/DC
converter controller
forms a typical synchronous-rectified step-down converter
(Figure 1).

Figure
1. This illustrative step-down switching regulator features an
externalswitching transistor (N1) and synchronous rectifier
(N2).

During normal operation, the circuit conducts
current from input to output when the high-side
switch (N1) is on, and it continues conducting through the
inductor when N1 is off and the synchronous rectifier (N2) is
on. First-order approximations of the current and voltage
waveforms (Figure 2) lead to a flawed assumption that
all the components are ideal, but the parasitic effects of
these components will be covered later.

Figure
2. These waveforms from the circuit in Figure 1 are based on
an assumption of ideal components.

Because N1 is on
only part-time, the input source and input capacitor
(CIN) see discontinuous currents. CIN
supplies the excess current (IL -IINPUT)
while N1 is on, and it stores charge from the input current
while N1 is off. If CIN were of infinite value,
with zero equivalent
series resistance (ESR) and equivalent
series inductance (ESL), the voltage across it would
remain constant during these partial charge and discharge cycles.
Actual voltage fluctuates over each cycle, of course. The
current pulses divide between CIN and the input
source, based on the relative conductance at or above the
converter's switching frequency.

One way to eliminate
these conducted emissions is the brute force approach: Connect
low-impedance bypass capacitors at the input. Yet a more
subtle approach can save cost and board area: Add impedance
between the source and the converter, making sure the
necessary DC current can pass. The best impedance is an
inductor, but take care that the converter's input impedance
remains low up to its loop crossover
frequency. (The loop crossover for most DC/DC switching
converters is between 10kHz and 100kHz.) Otherwise,
input-voltage fluctuations can destabilize the output voltage.

Current ripple on the output capacitor
(COUT) is much less than on CIN. Its
amplitude is lower, and (unlike the input capacitor) its
current is continuous, and therefore has less harmonic
content. Normally, each turn of the coil is covered with wire
insulation, forming a small capacitor between each pair of
turns. Adding these parasitic capacitors in series forms a
small equivalent capacitor in parallel
with the inductor, which provides a path for conduction of
current impulses to COUT and the load. Thus, the
discontinuous edges of the voltage waveform at the switching
node (LX) conduct high-frequency current to COUT
and the load. The usual result is spikes on the output
voltage, with energy in the 20MHz to 50MHz range.

Often the load for this type of converter is some form
of microelectronics susceptible to conducted noise, and
fortunately the converter's conducted noise is easier to
control at the output than at the input. As for the input,
output conducted noise can be controlled by very low impedance
bypassing or secondary filtering. One should be cautious of
secondary (post) filtering, however. Output voltage is a
regulated variable in the control loop, so an output filter
adds delay or phase (or both) to the loop gain, possibly
destabilizing the circuit. If a high-Q LC post-filter is
placed after the feedback point, the inductor's resistance
will degrade the load regulation, and transient load currents
can cause ringing.

Other Topologies
Other switching-converter topologies
have problems similar to those of the step-down converter. A
step-up converter (Figure 3), for example, has the
basic structure of a step-down converter, but with inputs and
outputs swapped. Thus, problems at the input of a step-down
converter apply to the output of a step-up converter as well,
and vice versa.

Figure
3. This step-up switching regulator lacks synchronous
rectification, but is otherwise similar to the step-down type,
with inputs and outputs swapped.

Step-down
converters are limited, because their output voltage must be
less than the input voltage. Similarly, a step-up converter's
output voltage must be greater than its input voltage. This
requirement is problematic when the output voltage falls
within the input-voltage range. A topology that solves this
problem is the flyback converter (Figure
4).

Figure
4. A flyback regulator maintains regulation for inputs that
range above and below the output voltage.

Because
currents at the input and the output are both discontinuous,
making conducted emissions more difficult to control, noise
from this converter is generally worse than that of a step-up
or step-down type. Another problem with this converter is that
current in each transformer winding is discontinuous, and
these discontinuities act with the transformer's leakage
inductance to produce high-frequency spikes, which can
conduct to other circuits. Physical separation of the primary
and secondary windings causes this leakage inductance. Thus,
the leakage inductance results from magnetic fields in the air
(because fields in the core couple both the primary and
secondary windings). Spikes due to the leakage inductance
therefore cause magnetic field radiation.

Another
topology that solves the problem of overlapping input and
output voltages is the single-ended primary inductance
converter (SEPIC). Similar to a flyback circuit, the SEPIC
converter connects a capacitor between the transformer's
primary and secondary windings (Figure 5). This
capacitor, which provides a path for current in the primary
and secondary windings during the time that flyback currents
are off, improves the flyback circuit by making the primary
and secondary currents continuous. On the other hand, adding
input or output capacitance to a flyback circuit can often
improve its emissions sufficiently to make that topology just
as acceptable. If conducted and radiated noise is expected to
be a problem, however, the SEPIC circuit can be preferable to
the flyback.

Figure
5. Otherwise similar to a flyback regulator, the single-ended
primary inductance converter (SEPIC) has continuous primary
and secondary currents that generate less noise.

Linear Post-Regulation
For some applications in which
output noise must be minimized, the efficiency deficit of
using a linear regulator is not acceptable. A switching
regulator followed by a linear post-regulator can be suitable
in these cases. The post-regulator attenuates high-frequency
noise generated by the switching regulator, resulting in noise
performance approaching that of a lone linear regulator.
Because most voltage conversion occurs in the switching
regulator, however, the efficiency penalty is much smaller
than for a lone linear regulator.

This scheme can also
replace flyback and SEPIC converters in applications for which
the input and output voltages overlap. The step-up converter
operates when the input is less than the output, and the
linear regulator operates when the input is greater than the
output. A step-up converter and a low
dropout (LDO) linear regulator can be combined in a single
IC (Figure 6). This device also includes a track mode
in which the step-up converter output voltage is always 300mV
above the LDO output voltage. As a result, the LDO regulator
maintains sufficient PSRR
and headroom (input minus output voltage) to attenuate noise
from the step-up converter under all conditions.

Figure
6. As a third option for maintaining regulation when the input
range overlaps the output voltage, this IC combines a
switching regulator (for step up) and a linear regulator (for
step down).

Common-Mode Noise
By definition, common-mode
conduction is in phase on both connections of the input or the
output. Typically, it poses a problem only for fixed systems
that have a path to earth ground. In a typical offline power
supply with common-mode filters (Figure 7), the main
source of common-mode noise is the MOSFET. The MOSFET is
usually a major power-dissipating element in the circuit, and
it requires a heatsink.

Figure
7. Common-mode filters in this typical offline power supply
reduce noise that is common to both sides of the input and the
output.

For a TO-220 device, the heatsink tab would
connect to the MOSFET drain,
and in most cases the heatsink would conduct current to earth
ground. Because the MOSFET is both insulated and electrically
isolated from the heatsink, it has some capacitance to earth
ground. As it switches on and off, the rapidly changing drain
voltage drives current through the parasitic capacitance
(CP1) to earth ground. Because the AC line has low
impedance to earth ground, these common-mode currents flow
from the AC input to earth ground. The transformer, too,
conducts high-frequency current through the parasitic
capacitance (CP2A and CP2B) between its
isolated primary and secondary windings. Thus, noise can be
conducted to the output as well as the input.

In Figure
7, the common-mode conducted noise is attenuated by the
common-mode low-pass filters between the noise source (the
power supply) and the input or the output. Common-mode chokes
(CML1 and CML2) are generally wound on a single core with the
polarity shown. Load current and the line current driving the
power supply are both differential-mode currents (that is,
current flowing in one line flows out the other). By winding
the common-mode chokes on a single core, the fields due to
differential-mode currents cancel, allowing use of a smaller
core because very little energy is stored in it.

Many
of the common-mode chokes designed for offline power supplies
are wound with physical separation between the windings. This
construction adds differential-mode inductance, which also
helps to reduce the conducted differential-mode noise. Because
the core links both windings, fields due to differential-mode
current and inductance are in the air rather than the core,
which can produce radiated emissions.

Common-mode noise
generated in the power supply's load can be conducted through
the power supply to the AC line by means of parasitic
capacitance (CP2A and CP2B) in the
transformer. A Faraday shield in the transformer (a ground
plane between primary and secondary) can reduce this noise
(Figure 8). The shield forms capacitors from the
primary and secondary to ground, and these capacitors shunt
common-mode currents to ground rather than allowing them to
pass through the transformer.

Figure
8. A Faraday shield between primary and secondary blocks
common-mode noise that would otherwisepass through the
transformer's parasitic interwinding
capacitance.

Just as conducted emissions can be in
the form of voltage or current, radiated emissions can be in
the form of electric or magnetic fields. Because fields exist
in space rather than in conductors, however, there is no
distinction between differential-mode and common-mode fields.
An electric field exists in the space between two potentials,
and a magnetic field exists around a current traveling through
space. Both fields can exist in a circuit, because capacitors
store energy in electric fields and inductors/transformers
store/couple energy in magnetic fields.

Electric Fields
Because an electric field exists
between two surfaces or volumes with different potentials, it
is relatively easy to contain the electric field noise
generated within a device by surrounding the device with a
ground shield. Such shielding is common practice in the
construction of CRTs, oscilloscopes, switching power supplies,
and other devices with fluctuating high voltages. Another
common practice is the use of ground planes on circuit boards.
Electric fields are proportional to the potential difference
between surfaces and inversely proportional to the distance
between them. They exist, for instance, between a source and
any nearby ground plane. Multilayer circuit boards therefore
let you shield circuitry or traces by placing a ground plane
between them and any large potential.

Yet one should be
cautious of capacitive loading on high-voltage lines when
using ground planes. Capacitors store energy in electric
fields, so placing the ground plane near a conductor forms a
capacitor between the conductor and ground. A large dV/dt
signal on the conductor can cause large conducted currents to
ground, thereby degrading the conducted emissions while
controlling the radiated emissions.

If electric field
emissions are present, the most likely culprit is the highest
potential in the system. In power supplies and switching
regulators, beware of the switching transistors and
rectifiers, because they normally have high potentials and can
also have large surface areas due to heatsinking.
Surface-mount devices can have this problem, too, as they
often require lots of circuit-board copper for heatsinking. In
that case, also watch out for capacitance between any
large-area heatsink plane and the ground plane or a
power-supply plane.

Magnetic Fields
Electric fields are relatively easy to
contain, but magnetic fields are a different proposition.
Enclosing a circuit in high-µ material can provide an
effective shield, but that approach is difficult and costly.
Usually the best way to control magnetic field emissions is to
minimize them at the source. In general, this requires that
you choose inductors and transformers designed to minimize
radiated magnetic fields. What is equally important is for the
circuit-board layout and interconnect wiring to be configured
to minimize the size of current loops, especially in
high-current paths. Not only do high-current loops radiate
magnetic fields, but they also increase the inductance of
conductors, which can cause voltage spikes on lines that carry
high-frequency current.

Inductors
Circuit designers not experienced in
transformer or inductor design are likely to choose
off-the-shelf transformers and inductors. Even so, a bit of
knowledge about magnetics can enable a designer to select the
optimal components for an application.

The key to
reducing inductor emissions is using high-µ material for
keeping the field in the core and out of the surrounding
space. Magnetic fields have a proportionally higher density in
higher-µ material. This is much like parallel conductance: A
1S conductance (i.e., a 1Ω resistor) in parallel with a 1mS
conductor (1kΩ resistor) has 1000 times' the current of the
1mS conductor. A magnetic field density divides in a ratio of
1000:1 between a 1000µ, 1in² core and a 1µ, 1in² core. High-µ
materials cannot store a lot of energy, so for compact
inductors one must employ a high-µ core with an air
gap.

To understand why, look at Figure 9. The
B-field (X-axis) is proportional to V×t/N, where N is the
number of turns. The H-field (Y-axis) is proportional to N×i.
Thus, the slope of the curve (proportional to µ) is also
proportional to the inductance (L = V/[di/dt]). Adding a gap
to this ferrite (or any other high-µ core) reduces the slope,
thereby lowering the effective µ and consequently the
inductance. Inductance decreases by the change in slope,
maximum current increases by the change in slope, and the
saturation B-field remains the same. Therefore, the maximum
energy (½LI²) stored in the inductor increases. This increase
can also be illustrated by applying a voltage to the inductor
and noting the amount of time to reach Bsat. Energy stored in
the core is the integral of (V×i)dt. Because the current
associated with a gapped core is higher for the same voltage
and time, the corresponding level of stored energy is
higher.

Figure
9. Gapping a ferrite core forces magnetic flux out of the core
and allows the inductor or transformer to store energy in a
field surrounding the device.

Gapping the core,
however, increases magnetic field radiation in the space
around the inductor. Bobbin cores, for example, whose large
air gap makes them notorious generators of magnetic field
radiation, are generally avoided in some noise-sensitive
applications for that reason. The bobbin core, which is just a
bobbin-shaped piece of ferrite, is one of the simplest and
cheapest types of gapped ferrite core. Wire is wound around
the center post to make an inductor. Costs are low because the
wire can be wound directly around the core with no extra work
other than terminating the wire. In some cases the wires are
terminated on a metallized area at the bottom of the core,
allowing the inductor to be surface-mounted. In other
surface-mounted components, the inductor is mounted on a
ceramic or plastic header, to which the wires are
terminated.

Some manufacturers put ferrite shields
around the bobbin core to help reduce field emissions. This
measure helps, but it also reduces the gap and therefore
decreases the energy that can be stored in the core. Because
the ferrite itself can store very little energy, a small gap
is often retained between the shield and the core, which
allows some unwanted radiation of magnetic fields in this type
of inductor. Depending on the level of acceptable emissions,
however, the bobbin core can be a good compromise between cost
and EMI.

Various other core shapes can be gapped (or
not) according to the requirements of the application. Pot
cores, E-I cores, and E-E cores, for example, all have center
legs or posts that can be gapped (Figure 10). Gapping
the center of the core, which is completely surrounded by the
coil, helps to reduce the emissions radiated from the air gap.
These inductors are usually more expensive, because the coil
must be wound separately from the core, and the core parts
assembled around the coil. For easy design and assembly, cores
can be purchased with a pregapped center leg.

Figure
10. Different core geometries offer trade-off among energy
storage, field emission, and ease of assembly. All can be
gapped.

Perhaps the best core for reducing radiated
emissions is the distributed-gap toroid. This core is made by
pressing a mixture of filler and high-µ metal powder into the
doughnut shape of a toroid. The grains of metal powder,
separated by nonmagnetic filler, have small air gaps between
them that create an overall "air gap" distributed evenly
throughout the core. The coil is wound through the center and
around the outside of this core, making the field travel in a
circle along the middle of the coil. As long as the coil is
wound around the whole circumference of the toroid, it shields
the outside by surrounding the magnetic field
completely.

The loss in a typical distributed-gap
toroidal core is sometimes higher than for gapped ferrites,
because metal grains in the toroid are susceptible to eddy
currents that generate heat and reduce power-supply
efficiency. Toroids are also expensive to wind, because wire
must be fed through the center of the core. Machines can do
this, but they are slower and more expensive than traditional
coil-winding machines.

Some ferrite toroidal cores
have a discrete air gap. The resulting magnetic field
emissions are higher than those of distributed-gap cores. Yet
typical gapped toroids have lower losses, because they contain
the field better than other discretely gapped ferrite cores.
The coil reduces emission by shielding the gap, and the
toroidal shape helps to keep the field inside the core.

Transformers
Transformers have many limitations in
common with inductors, because they are wound on the same
cores. Some issues are unique to transformers, however. The
performance of actual transformers can approach that of an
ideal transformer— coupling voltage from primary to secondary
with a ratio of voltages proportional to the ratio of turns in
each winding.

The equivalent circuit of a transformer
(Figure 11) models the interwinding capacitance as
CWA and CWB. The problem posed by these
parameters is mainly that of common-mode emissions in isolated
power supplies. Winding capacitances CP and
CS are small and usually negligible at the
operating frequencies of switching power supplies and
regulators. The magnetizing inductance LM is important,
because too much magnetizing current can cause the transformer
to saturate. As for inductors, saturation increases the
magnetic field emission from the transformers. Saturation also
causes higher core loss, higher temperature
(with the possibility of thermal runaway), and a degradation
of coupling between the windings.

Figure
11. Parasitic elements in the equivalent model for a
transformer modify its ideal behavior.

Leakage
inductance is caused by a magnetic field that links one
winding but not the other. Though some coupled inductors and
transformers (like the common-mode choke discussed earlier)
are designed for a high level of this parameter, leakage
inductances LLP and LLS are the most
problematic parasitic elements in a switching power supply.
Magnetic flux that links two windings couples those windings
together. All transformer windings are around the core, so any
leakage inductance is outside the core, in the air, where it
can cause magnetic field emissions.

Another problem
with leakage inductance is the large voltage generated when
the current changes quickly, as it does in the transformer of
most switching power supplies. Such voltage can overstress the
switching transistor or rectifier. Dissipative snubbers
(usually a series resistor and capacitor) are often used to
control this voltage by dissipating the energy of the voltage
spike. On the other hand, some switching devices are designed
to withstand repetitive avalanche breakdowns and can dissipate
the energy without external snubbers.

The leakage
inductance of a transformer can be determined by shorting the
secondary and measuring the inductance at the primary. This
measurement includes any secondary leakage inductance coupled
through the transformer, but in most cases such leakage must
be accounted for anyway because it adds to the primary voltage
spike. The corresponding spike energy is calculated as E =
½LI², so power lost to the leakage inductance is the energy of
each spike multiplied by the switching frequency: P =
½LI²f.

Transformer requirements depend on the
power-supply topology. Topologies that couple energy directly
across the transformer, such as half-bridge, full-bridge, push-pull,
or forward converters, require a very high magnetizing
inductance to prevent saturation. The transformer's primary
and secondary simultaneously conduct current in these
circuits, coupling the energy directly through the
transformer. Because little energy is stored in the core, the
transformer can be smaller. These transformers are wound
typically on an ungapped core of ferrite or other high-µ
material.

Other power-supply topologies require that
the transformer core store energy. The transformer in a
flyback circuit stores energy by means of the primary in the
first half of the switching cycle. In the second half of the
cycle, energy is retrieved and fed to the output by means of
the secondary. As is true for inductors, an ungapped high-µ
core is not suitable for storing energy in a transformer.
Instead, the core must be gapped discretely or have a
distributed gap. The resulting component will be larger than
one with an equivalent ungapped core, but it can save an extra
inductor as well as cost and space.

Layout
Component selection is very important in
controlling EMI, but the circuit-board layout and
interconnects are equally important. Especially for the
high-density, multilayer circuit boards often used in
switching power supplies, layout and component placement are
critical to the circuit's proper operation and interaction.
The power switching can cause large dV/dt and di/dt signals in
the circuit-board traces, which lead to compatibility problems
by coupling to other traces. Compatibility problems and
expensive circuit-board revisions can be avoided, however, by
taking extra care in the layout of critical paths.

A
distinction can be made between radiated and conducted
emissions in a system, but the distinction blurs when talking
about interference in a circuit
board and wiring. Adjacent traces that couple electric
fields also conduct currents by means of parasitic
capacitance. Likewise, traces that are coupled by magnetic
fields act somewhat like transformers. These interactions can
be described in terms of lumped components or through field
theory. Which approach to take depends on which method more
accurately describes the interaction.

Crosstalk
Two or more conductors in close proximity
are coupled capacitively; therefore, large voltage changes on
one will couple currents to the other. If the conductor's
impedance is low, the coupled currents generate only small
voltages. Capacitance is inversely proportional to the
distance between the conductors and proportional to the area
of the conductors, so the conducted noise can be minimized by
keeping the area of the adjacent conductors small and their
separation large.

Another method for reducing the
coupling between conductors is to add a ground plane or
shield. A ground trace (or in some cases a power-supply bus or
other low-impedance node) between conductors can prevent their
interaction by coupling them capacitively to ground instead of
to each other. But exercise caution. Traces carrying fast
dV/dt changes, positioned close to a plane with high-impedance
interconnect to ground, can couple these changes to the ground
plane. In turn, the ground plane can couple the signals to
sensitive lines, thereby exacerbating rather than helping the
noise problem. If the ground plane doesn't carry large
currents, it can be tempting to connect it to ground through a
small wire. Yet the high inductance of a small wire can cause
the ground plane to look like a high impedance to
fast-changing voltages.

One must take care to ensure
that a ground plane does not inject noise into sensitive parts
of a circuit. Input and output bypass capacitors, for example,
often pass current through a ground plane, and the
high-frequency current components can affect sensitive
circuitry. To prevent this problem, circuit boards often
include separate planes for the power and signal grounds.
Connected at a single point, these planes minimize the noise
injected into signal ground by potentials generated across the
power-ground plane. This practice is similar to that of a star
ground, in which all components connect to ground at a
single point (all traces leave that point in a "star"
pattern). The star ground has the same effect as separate
power- and signal-ground planes, but it isn't practical for
large, complex circuit boards including lots of grounded
components.

If a node is known to be sensitive to
injected noise, then traces and wires connected to that node
should be routed away from nodes with high-voltage changes. If
that isn't possible, add a good ground or a shield. Good
capacitive bypassing of the node can also decrease its
susceptibility to crosstalk.
Normally, a small capacitor connected between the node and
ground or between the node and a power-supply bus forms a
suitable bypass.

When choosing the bypass capacitor,
make sure it has low impedance over the range of frequencies
that are problematic potentially. Equivalent series resistance
(ESR) and equivalent series inductance (ESL) can cause the
impedance to be higher than expected at high frequencies, so
the low ESR and ESL of ceramic capacitors is attractive for
bypass applications. The ceramic dielectric has a major effect
on performance as well. Higher-capacitance dielectrics (such
as Y5V) can allow large changes in capacitance over voltage
and temperature. At maximum-rated voltage, capacitors made
with these ceramics can exhibit as little as 15% of their
unbiased capacitance. A smaller capacitance value with a
better dielectric, yielding crosstalk attenuation that is not
as dependent on bias and temperature, will in many cases
provide better and more consistent bypassing.

The
placement of bypass capacitors is also critical. To attenuate
high-frequency noise, you must route the signals in question
through the bypass capacitor. In Figure 12a, the length
of trace in series with the capacitor adds to its ESR and ESL,
increasing the impedance at high frequencies and reducing the
capacitor's effectiveness as a high-frequency bypass. A better
layout routes the trace through the capacitor, so the traces'
stray ESR and ESL aid the bypass capacitor's filter action
rather than degrading it.

Figure
12. Poor bypass connections (a) add trace inductance and
resistance to the capacitor. In the better connection (b),
trace parasitics add to the capacitor's filter
action.

Some nodes should not be bypassed, because
doing so changes their frequency characteristics. An example
is the feedback resistor divider. In most switching power
supplies, a resistive feedback divider drops the output
voltage down to a level acceptable to the error amplifier.
A large bypass capacitor added to this feedback node forms a
pole with the resistance of that node. Because the divider is
part of the control loop, this pole becomes part of the loop
characteristic. If the pole frequency is less than one decade
above the crossover frequency, its phase or gain effects can
affect the loop stability adversely.

Inductance
Some currents in a switching power supply
switch on and off quickly. Stray inductance in those current
paths can induce large noise voltages, which couple into
sensitive circuitry and stress the components. Lines carrying
DC currents seldom lead to problems, because DC does not cause
voltage spikes or couple AC to other traces. A line in series
with an inductor, for instance, is not a problem, because the
stray inductance is much smaller than the inductor value. The
large series inductance also prevents discontinuities in the
current.

If a circuit produces discontinuous currents,
try to prevent the current from traveling in large loops.
Large loops of current produce larger values of inductance,
thereby increasing any consequent magnetic field radiation.
This caveat applies to component placements as well, because
current usually switches between active devices such as
transistors and diodes.

Consider the step-down
converter in Figure 1. When the high-side MOSFET switch (N1)
is on, current travels by means of the input, N1, the
inductor, and the load. After N1 turns off, the diode
(D) conducts current until the synchronous rectifier (N2)
turns on. Current then flows through N2 until it turns off;
then the diode carries current until the cycle starts again.
Note that currents through the inductor and the output
capacitor are continuous, and therefore should not be major
contributors to noise. If N1, N2, and D are placed some
distance from one another, the surrounding fields must shift
quickly, in response to the rapid current changes within them.
Because the voltage generated is proportional to the change in
the magnetic field with time (dΨ/dt), these rapid field
fluctuations can generate large voltage spikes.

Note
that the input source and the output load carry high-frequency
currents. These currents should pass through the input and
output bypass capacitors; otherwise, they are conducted
through the input or output lines (or both). See the section
on conducted emissions. The impedance of input and output
bypass capacitors is important. They should be large enough to
keep the impedance low at input and output, but larger
capacitors (tantalum or electrolytic types, for instance) have
higher ESR and ESL than smaller ceramic types. You must
therefore ensure that the impedance of the capacitors is
sufficiently low at the frequencies of concern.

One
alternative is to parallel a ceramic capacitor with an
electrolytic or tantalum capacitor, because the ceramic
capacitor has lower impedance at high frequencies. In most
cases, however, that arrangement is no better than multiple
electrolytic or tantalum capacitors in parallel to reduce ESR
and ESL, or multiple ceramic capacitors in parallel to
increase the total capacitance.

A similar article
appeared in the October 26, 2000 issue of EDN.

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APP 716: Aug 21, 2002

AN716, AN 716, APP716,
Appnote716, Appnote 716

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