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John
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30 November 2014

OpAmps as Input Stages

OpAmp-Based Input Stages

IC OpAmps make an electrical engineer's job easier to
do, as they do the work of a hundred discrete components, yet are no
bigger than a small spider and vastly smaller, for example, than a 1µF PIO
coupling capacitor. And while the first OpAmps, such as the LM741, sounded
just dreadful, the newer OpAmps, such as the OPA637, sound quite good. For
us tube-loving folk, however, one big problem presents itself when dealing
with OpAmps in a tube projects: their low maximum power-supply voltage
limits, seldom more than 36Vdc from negative power-supply pin to positive
pin.

Well, a few companies actually do make high-voltage
OpAmps, such as Apex;
and we can overcome this limitation with some clever design. Nonetheless,
let's start with a simple—albeit a bit unrealistic—example: a two-stage,
hybrid, single-ended power amplifier. Unrealistic? The OpAmp in this
example sees a differential of 185 volts from its negative power-supply
pin to its positive power-supply pin, which certainly counts as
unrelealistic. Let's ignore this, however, and examine the rest of the
circuit.

The input is capacitor coupled and a +10V bias voltage
is applied at the OpAmp's inverting input via the 100k resistor. Due to
the phase inversion produced by the 300B output tube being configured as a
grounded-cathode amplifier, the OpAmp's inverting input is used; thus,
effectively the OpAmp's inverting input becomes a non-inverting input,
while the OpAmp's normally non-inverting input becomes the inverting
input. The OpAmp's output DC couples to the 300B's grid via the 10k
grid-stopper resistor. From the 300B's plate, a two-resistor negative
feedback extends down to ground.

The negative feedback loop works to establish an equal
voltage on the OpAmp's two inputs at all times. This results in a fixed
gain and low output impedance at the 300B's plate. Moreover, it greatly
reduces the distortion at the plate, as the plate is the now effectively
the output of this large hybrid OpAmp. In other words, the negative
feedback loop does not include the output transformer. This is an
important detail, as we will see later on. But for right now, we are in
Happy Land; thus the happy face.

One feature about which to be happy is the auto-biasing
of the output tube, which results from the negative feedback and the 10V
of bias voltage at the OpAmp's input. Once again, the OpAmp does all it
can to maintain the exact same voltage at each of its two inputs, which
means that it will try to force the 300B's plate voltage to exactly
400Vdc, as the two-resistor voltage divider that constitutes its negative
feedback from the plate can only establish +10V with a plate voltage of
400Vdc. Since the regulated B+ connection delivers 410Vdc and since the
output transformer primary's DCR (DC resistance) equals 133 ohms, the
current flow through the DCR must equal 75mA, as 10V/133-ohms equals 75mA.
In other words, auto-bias. As the output tube ages or is replaced, the
same 75mA of idle current will obtain.

Like Disneyland and Fantasy Island, Happy Land is a
great place to visit, but no really lives there. Here in Reality Land,
things get tough, plenty fast. For example, very few power amplifiers,
either solid-state or tube-based, enjoy a regulated B+ for the output
devices. Sure, some are lucky enough to use high-voltage regulators for
the input and driver stages, while others use some form of AC-only
regulation on the B+ connection to the output stage, wherein ripple is
scrubbed away, but no attempt is made to maintain a fixed DC output
voltage. Why not? Harsh reality.

The wall socket does not put out a fixed AC voltage, so
a regulator must be able to contend with brown-outs and over voltages.
Let's be forgiving and allow for only 10% slumps and 10% swells of the
nominal wall voltage. Thus, the raw DC power supply behind the
high-voltage regulator would see its raw DC fall by 10% or rise by 10%.
Now, let's assume that a 410V regulator requires 40V of headroom, so the
raw DC power supply voltage would be 450Vdc. If we reduce this voltage by
10%, we end up with 405Vdc, which, obviously enough, is not enough to
deliver the required 410Vdc of output voltage. If we increase the 450Vdc
voltage by 10%, we end up with 495Vdc, which means that the high-voltage
pass device would have to handle the increased dissipation due to the more
than doubling of the voltage drop across the pass device. Yet, in order to
overcome the 10% droops in wall voltage, we will have to run a nominal
500Vdc of raw power supply voltage, 100%/90% against 450V equals 500Vdc.
In other words, with a 90% of the normal wall voltage available, we still
get 450Vdc of raw power supply voltage.

And we have just been warming up, as not only is the
wall voltage poorly regulated, the power transformer adds its poor
regulation, which with high-voltage transformers can easily be 25%. What
is transformer regulation? It is the measure of the ratio between the
secondary's AC output voltage with and without a load. Some low voltage
toroidal power transformer run 2% to 5% regulation figures, but most
high-voltage power transformers, particularly those that use I & E
laminations, run regulation figures closer to 20% to 40%. Assuming just a
regulation figure of 25%, the 500V power transformer will put out 625Vdc
with no load, such as when the tubes are cold and not conducting, which
means that we can forget using 560V power-supply capacitors. In fact, we
must account for a possible 10% over-voltage from the wall socket, so the
625Vdc can become 687Vdc.

So much for breezily regulating the B+ voltage for the
output stage of a tube power amplifier. In other words, the 410Vdc will
not be regulated in Reality Land. The result is that the nominal 410V B+
voltage will vary with the wall voltage. Is this a big deal? Sure it is.

Here is small example: say the wall voltage climbs by
only 5%, which it can easily do by just moving to another part of your
town or from moving from the ground floor to the 20th floor of an old
apartment building. The B+ voltage climbs to 430Vdc, just 20V more, but
the output tube dissipation goes up by threefold, from 30W to 90W.

In addition, the unregulated power supply will not
offer the regulator's clean, ripple-free voltage, which will become a much
bigger problem due to the negative feedback loop. The OpAmp will strive to
keep the output at the plate absolutely noise free, so the output
transformer primary will see no ripple at the plate end and all the ripple
at the B+ connection. Not good. The output transformer will treat the
ripple as signal that must be transferred to the secondary and your
loudspeaker. Now, you can see why so many prefer virtual reality to real
reality.

But do not despair; like any good magician, I have many
tricks up my sleeve.

What we need to do is find a way to measure the DC
voltage drop across the output transformer primary at idle, which means we
have to inform the OpAmp what the B+ voltage is.

Below, we see a pair of two-resistor voltage dividers
and an added 10V zener diode. The zener diode creates a reference voltage
that the OpAmp uses to maintain fixed 10V voltage drop across the
transformer primary, as that is the only way that the OpAmp can see equal
voltages on its two inputs.

Well, how well does this auto-bias trick work at
eliminating power-supply noise from the output? Extremely well. Think
about it: the power-supply noise is just fast moving B+ voltage
variations. Now it might seem paradoxical that we want the 300B plate to
see all of the power-supply noise that the B+ holds, but that is only
because we tend to think only in terms of ground reference. But from the
perspective of the output transformer primary, if it sees no delta, no
difference between its leads, then it will ignore whatever common-mode
signal is present. Another way of putting it is that the transformer's
primary is B+ referenced, not ground referenced.

Okay, now that we have taken our baby steps, we can
make the adult leap forward to an actually usable power amplifier.

We have given the OpAmp +/-15V power-supply rails and
added a high-voltage PNP transistor, an MJE350. Because the transistor
inverts its input signal, we must invert the OpAmp inputs, so that the
inverting input is used as the inverting input for the entire power
amplifier. The amplifier's input signal is capacitor coupled with the 1µF
capacitor and the two-resistor voltage divider feeds the slightly
attenuated input signal to the OpAmp's non-inverting input.

The assumption here is that the input signal source
will offer a low input impedance, so that the two-resistor voltage divider
can do its magic. If a 100k volume potentiometer is used in front of the
coupling capacitor, then the deal is off, as the potentiometer's high
output impedance, as much as 25k with a 100k potentiometer, will undo the
AC voltage division. (The DC voltage division will still obtain, as will
the auto-bias functionality.) If a volume potentiometer or high-impedance
signal source must be used, then the following workaround will offer the
required low input impedance for the two-resistor voltage divider.

The added 1N4148 signal diode was needed to compensate
for the 0.6V voltage drop from the transistor's base to its emitter. This
hybrid power amplifier uses a negative power-supply rail, which we may not
have readily available, so the following design uses a monopole
high-voltage power supply.

The B+ voltage has been upped to 560Vdc, which is a
convenient voltage, as 400Vac rectifies up to 560Vdc. The 150V zener
establishes the DC voltage window within which the NPN transistor works.
This is still a fixed bias arrangement, as the KT120's grid attaches to
the (110V) collector, not a grid resistor to ground. The KT120 is operated
as a pentode, which would normally present a high plate impedance; but
with the negative feedback loop in place, the output impedance at the
plate will be plenty low.

Once again, the auto-bias feature ensures that at idle,
10V of voltage will drop across the output transformer primary. The
feedback resistors set a gain of (545k + 5k)/5k or 110, which means that
about 3Vpk of input signal will be needed to bring the amplifier to full
output. (Remember that the input two-resistor voltage divider's slight
attenuation must be factored in the equations.)

As I came up with this design, my first impression was
that this would make a fine music power amplifier. This then led to my
thinking that an extra OpAmp would not be out of place in such an
amplifiers either a guitar or microphone input would require lots more
gain, which an extra OpAmp could easily provide.

Note how we are deriving the OpAmp power supply from
the 20V zener diode. By using two 15V zeners, we could have created an
artificial ground for the OpAmps and given the OpAmps +/-15Vdc
power-supply rails. But I think that I will stop right here, as there is
plenty to feast on already.

Next Time
Details
about some new PCBs.

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//JRB

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