electron's

OPTIMISED ELECTRON
STREAM © TECHNOLOGY

in

AUDIO and RF
MODULATOR

TRIODE, TETRODE, PENTODE AND BEAM POWER
TUBES

Readers, please note this
page is presented for your education, information and guidance only - it is not
intended to be a technical/scientific treatise. The concepts and ideas presented
herein are just as much subjective as objective.

It attempts to assist
those experienced audiophile home constructors who wish to
explore further tube amplifier development options and are prepared "to go
beyond the square" - to challenge the paradigm of "the
status-quo".

This paper refers only to
the characteristics and performance of push-pull tube audio amplifiers without
negative feedback.

For reasons detailed
elsewhere in my website I have no interest whatsoever in single-ended
amplifiers - however the concepts described herein are as equally applicable to
single-ended amplifiers as to push-pull configurations.

For reasons detailed
elsewhere in my website I have no interest whatsoever in the use of trans-stage
negative feedback thus the concepts described herein are intended to optimise
electron tube performance without the use of negative feedback.

For full ratings and
applications of specific tube types in which you are interested please refer to
the manufacturer's catalogue.

Copyright in all quoted
works remains with their owner, author and publisher, as
applicable.

Please note that no
warranty is expressed or implied - see footnote notice.

Excepting where rights
in Intellectual Property previously exist, all rights in the applied engineering
concepts expressed in this paper remains exclusively with the
author.

The term OPTIMISED ELECTRON STREAM © TECHNOLOGY applies
to all part or whole uses of the concept.

The whole or part
thereof of this paper and/or the designs and design concepts expressed therein
may be reproduced for personal use only without limitation - but must not under any circumstances be
applied or used for commercial gain or reward without the express written
permission of the author, being the intellectual copyright
owner.

All rights
reserved.

OPTIMISED ELECTRON STREAM © TECHNOLOGY

1.
INTRODUCTION

It is recommended that to
best understand the concepts presented here, the reader first carefully study
my explanatory papers.

To directly access please
click on each of the links below:

SCREEN-GRIDS

ULTRA-LINEAR

OPTIMISED
ULTRA-LINEAR © OPERATION

TRIODE
OPERATION OF TETRODES AND PENTODES

POWER
SUPPLIES

HOW
TO DESIGN AND CONSTRUCT A HI-FI TUBE AMPLIFIER

It is intended that the
set ofOPTIMISED ELECTRON STREAM ©
TECHNOLOGY concepts presented below represent -
individually or together as a set - an innovative extension of audio engineering
design thinking to the above linked presentations elsewhere on my website, and a
major breakthrough in electron power tube technology available to the home audio
constructor.

Historically we have been
restricted by convention to a small set of engineering design principles that
have imposed a barrier to further development, reinforced by commercial
manufacturers staying with the tried and true "safe" design
configurations.

However the availability
of low-cost high-performance solid state amplifiers and pre-amplifiers,
together with advances in recording formats - currently Super Audio CD and DVD
standards - have provided stimulus to tube hi-fi enthusiasts to improve the
performance of their existing equipment.

Many of the old
constraints are longer with us - hiss, noise, wow, flutter and rumble etc. do
not present on CD or DVD - so enable us to open the frequency range and dynamic
range window a little more.

Modern CD's and DVD's
offer a substantially higher dynamic range than traditional analogue vynil
recordings. This dynamic range is compressed when the amplifier is not capable
of reproducing it, resulting in a loss of fidelity and realism.

The OBJECT of
OPTIMISED ELECTRON STREAM © TECHNOLOGY is to OPTIMISE performance in a high-fidelity audio amplifier by
simultaneously loading ALL the internal elements of a power electron tube with a
load directly proportional to their ACTUAL applied voltages - AC and DC - as
distributed between the positive and negative terminals of the tube - ie anode
and cathode - AND in direct proportion to their physical electrode spacing
(dielectric strength).

Hence OPTIMISED ELECTRON STREAM © TECHNOLOGY
may also be described as OPTIMISED DISTRIBUTED
LOAD © TECHNOLOGY

OES  © TECHNOLOGY
offers superior performance because it is
simple and because the tube is always in equilibrium.

The conceptual model is
illustrated in the following circuit diagram for Beam Power Tubes, Tetrodes
and Pentodes in conventional or Ultra-linear mode  - as
applicable:

To prepare a foundation
for the concepts presented on this page, please read my SCREEN-GRIDS
page before reading this paper, because the function and behaviour of the
Screen-grid is fundamental to OPTIMISED
ELECTRON STREAM © TECHNOLOGY.

In the case of
ultra-linear connected TETRODES and BEAM POWER TUBES only the
Screen Grid is connected to the 40-50% turns transformer primary winding tap as
shown above (because there is no separate Suppressor Grid terminal on the
tube)

OPTIMISED ELECTRON STREAM © TECHNOLOGY: SCOPE

This paper introduces nine
new applied engineering OPTIMISED ELECTRON
STREAM © TECHNOLOGY concepts for refining and
optimising the electron flow in a vacuum tube and its control
circuitry.

Optimised
Screen-Grid Voltage and Load in Tetrodes, Pentodes and Beam Power
Tubes

Optimised Suppressor
Grid Voltage and Load in Pentodes

Silicon Diode Feed
in Triodes, Tetrodes, Pentodes and Beam Power Tubes

AC Circuit Bypass
Capacitors - Plate Circuit, Screen-grids, Suppressor-grids, Control-grids,
Cathode Bypass

RATIO of Screen-grid
and Suppressor Grid and Cathode Bypass to Plate Bypass
Capacitors

Single or double
stage PI filter in each power supply wherein the filter capacitors are of
equal capacitance value and each inductor filter choke is bypassed by a
reverse biased diode

Decoupling of the
Power Amplifier from the Power Supply by means of a series connected silicon
diode

Decoupling each
amplifier stage from the next following by means of a series connected silicon
diode

Symmetrical balanced
AC signal drive system whereby the central axis of the AC signal OUTPUT from
the Driver stage circuit and the AC signal INPUT of the Driven stage circuit
are at the same common reference voltage potential.

Subject to individual
circuit design, all or any of the above OPTIMISED ELECTRON STREAM © TECHNOLOGY concepts
may be used individually, or in combination, or all together - each offering a
specific benefit to enhancing electron flow within each tube or
circuit.

2.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:
OPTIMISED
SCREEN-GRID VOLTAGE
AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in "Ultra-linear", or
"Distributed Load" operation.
OPTIMISED
SCREEN-GRID VOLTAGE AND LOAD for Tetrodes, Pentodes
and Beam Power Tubes in "Pentode' operation
.
.

2.1     OPTIMISED SCREEN-GRID VOLTAGE AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in
"Ultra-linear", or "Distributed Load" operation.
.
My OPTIMISED
ULTRA LINEAR ©  page describes a design method for optimising the
DC supply voltage and load impedance to the Screen Grids in
Tetrodes, Pentodes and Beam Power Tubes for "Ultra-linear", or "Distributed
Load" operation.
.

To determine what optimum
transformer primary tap ratio to use it is essential to physically determine the
actual Screen Grid to Anode physical (linear distance) space ratio, because for
general audio class electron tubes the ratio of screen grid to plate voltage
will be optimally equal to the physical distance ratio of cathode to screen-grid
v cathode to plate.
.

.
2.2
OPTIMISED SCREEN-GRID
VOLTAGE AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in conventional
"Pentode' operation
.

The same principle for
optimisation of tube performance may be applied to conventional operation of
Tetrodes, Pentodes and Beam Power Tubes in "Pentode' operation - ie, where the
Screen-Grids are supplied from a separate DC source to the Plates.

.
To
optimise the Screen-Grid voltage, first carefully and accurately measure the
physical gap spacing between the Cathode, Screen-Grid and Plate.

In most cases this can be
easily done without having to destroy a valuable tube. Just measure the spacing
between Cathode and
Anode
and of the grid support pins at the top of the tube. In the case of beam power
tubes this may be difficult because there is
usually a box-like assembly covering the Plate structure. But a
bit of ingenuity should solve the problem

If it becomes necessary to
destroy a tube to examine its internals safety precautions must be
observed:

1.    Wrap
the tube in a strong cloth
2.    Place the tube in a vyce and squeeze the
glass bottle slowly until it implodes
or
3.
Gently hit the glass with a hammer at a point not directly over the electrode
assembly
4.    Carefully remove the broken glass and
dispose of safely
5.
Carefully cut away the electrodes until the elements can be
measured

The "gap" spacing ratio
can then be calculated for the particular tube type.

To determine the Optimised
Screen-Grid voltage, multiply the Plate voltage by the gap ratio.

It is obvious the
Screen-Grid voltage will be substantially less than the Plate voltage -
typically in the range 30 to 50 % but usually 40% - so a separate stable supply
is essential.

The designer has the
option of using:

a voltage divider
network from the Plate Supply (not recommended for
hi-fi)

a stabilised
Screen-Grid DC supply derived from the same AC source (power transformer) as
the Plate supply (not recommended for hi-fi)

a separate Screen-Grid
DC supply derived from a separate power transformer and full-wave rectifier
to the Plate supply (tube rectifier not recommended for
hi-fi)

a separate regulated
Screen-Grid DC supply derived from a separate power transformer and full-wave
rectifier to the Plate supply (tube rectifier not recommended for
hi-fi)

What is most important is for the Screen-Grid DC supply voltage
to be reasonably constant between no-signal load and full-signal load conditions
(ie well regulated).

This can be effectively
achieved by using a separate power transformer of generous rating - ie at least
twice the maximum signal DC Screen-Grid current, a full-wave silicon bridge
rectifier and humungous filter capacitors in capacitor input to filter
configuration - preferably with a low DC resistance filter choke installed
too.

For the more theoretical
designer some thought could be given to the "Virtual Cathode" concept, which
suggests that the portion of  electron stream devoted to the negative bias
applied to Grid #1 creates a more negative "Virtual Cathode" in the region of
the physical Grid element. This concept however is difficult to assess because
so long as the tube is conducting some of the electron stream is continuous
between real Cathode and Plate. Maybe only the outer portions of the electron
stream are affected by the negative bias and not the entire thickness of
stream.

The perfectionist - or
those having the necessary equipment - looking for the ultimate optimisation
could determine the optimum operating voltage for the Screen-Grids by means of
a Signal Generator, Distortion Meter and variable voltage power supply to the
Screen-Grids.

The optimised AC load
impedance for the Screen-grid will be the ratio of DC Screen-grid volts to Plate
volts x Plate load impedance.
If the Screen-grid load is provided
by a fixed resistor the ratio for a Screen-grid - or set of Screen-grids in one
side of a push-pull set, will be the DC voltage ratio x one quarter Plate to
Plate load impedance

Note however that if the
load is provided by a transformer as in ultra-linear or equivalent mode, the
Screen to Screen load impedance will be directly proportional, but the turns
ratio will be the square of the DC voltage ratio.

eg If the DC ratio is 40%
then the turns ratio for the Screen-grid tap on the output transformer will be
0.4 x 0.4 = 0.16 or 16%

A technical paper
describing benefits of reduced screen-grid voltage is presented at http://www.tubebooks.org/Books/lockhart.pdf

Recent Research

I am indebted to Rudolf  Moers, a
distinguished Electrical and Electronics Engineer located in the Netherlands,
who has made available for us his wonderful recent scientific investigation
into the design theory and practice of Ultra-linear audio amplification.

These papers are posted with permission from
Linear Audio www.linearaudio.net and their author Rudolf  Moers.

Part
1 - Power Point presentation

Part
2 - Power Point presentation

Parts
1 and 2 combined presentation - pdf format

Paper
- The Ultra-Linear Power Amplifier: An adventure between triode and pentode -
pdf format

The engineering design methods developed by Mr
Moers may be used to determine theoretical plate/screen load ratios for
ultra-linear operation of power tubes.

3.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:
OPTIMISED
SUPPRESSOR-GRID
VOLTAGE AND LOAD IN PENTODES

The PENTODE
electron tube has an extra grid (Grid #3) installed to suppress the effects of
secondary emission, as described in my SCREEN-GRIDS
page.

Although few true Pentodes are available for
audio power amplifier output stage applications, this section relates to them
because they are suitable for use in OPTIMISED ELECTRON
STREAM © TECHNOLOGY applications.

Manufacturers' tube data
sheets generally describe Pentodes as having higher measured distortion than
Beam Power Tubes so most audio amplifier designs have focused on the latter
types.

In the case of recycled or
salvaged tubes from yesteryear, it is evident that by WWII, RF design engineers
preferred Beam Power Tubes because of their capability to operate at higher RF
frequencies, hence the true Pentode enjoyed only a short claim to
fame.

One exception to this is
the mighty 803 Pentode, which found widespread use in submarine and
destroyer RF applications because of its ruggedness, stability, reliability and
long-life. Of interest is that the 803 pentode has aligned grids, each having a
ceramic coating to limit electron attraction, thereby improving predictability
and stability of operation, as well as efficiency.

The 837 Pentode
tube is also renown as a very reliable oscillator tube for RF transmitters. The
837 Pentode has a Plate structure similar to that of the famous TT21 and KT88
Beam Power Tubes.

However Pentodes do offer
user benefits over Tetrodes and have a place in high-fidelity audio and RF
amplifiers - particularly for home constructors who might have a box of useful
Pentodes just waiting to be used in a suitable design.

It is a well established tube engineering
principle that the current in a tube can be regulated using ANY of the grid
elements.

For example, in domestic radio receivers, the
use of multi-grid tubes such as 6A7, 6A8 and 6L7, each having 5 grids, is a
standard application of radio engineering design.

However for audio engineering purposes, three
grids appears to be the practical limit - beyond which no appreciable benefit is
to be realised. In fact. the beam Power Tube utilises only two grids to control
current through the tube.

Using the same applied engineering technology as
developed for OPTIMISED
ULTRA LINEAR © operation, it is possible and practicable in
power tubes fitted with a SUPPRESSOR-GRID having an independent base pin
connection, to apply a DC voltage and AC load impedance to the
SUPPRESSOR-GRID having the same ratio to Plate voltage and load
respectively, as the internal electrode physical gap ratio.

IMPORTANT NOTE 1: It is the case in many
of the popular PENTODES that the SUPPRESSOR-GRID is internally
connected during manufacture to the CATHODE, so it is not possible to
externally access the Suppresser Grid. In this class of tube the manufacturer
controls the behaviour of the Suppressor-grid and the user cannot do anything to
change that.

IMPORTANT NOTE 2: Tube identification
protocols describe and illustrate by base pinout diagrams Beam Power Tubes
as "Pentodes". These tubes are essentially Tetrodes that have a beam
forming electrode used to both confine the electron stream to a pre-determined
width within the tube, and to focus the electron flow onto a particular area of
the Plate. They usually have aligned Grids to arrange the electron flow in
multiple sheets.

Therefore it is essential to physically examine
tubes of interest to determine their actual mechanical construction.

DO NOT RELY ON DESCRIPTIONS OR BASE DIAGRAMS
PUBLISHED IN TUBE MANUFACTURERS' CATALOGUES, MANUALS OR DATA SHEETS.

However those tubes having
a separate base pin connection for the SUPPRESSOR-GRID offer the designer
the option of applying a suitable DC voltage consistent with the proportional
voltage divider effect of the internal electrode gap.

The rules and requirements
for an adequately regulated DC power source are the same as described above for
the Screen-Grid.

For example, in the Type
803 - 125W dissipation power transmitting pentode tube - the Suppressor
Grid is set at nominally 68% of the Cathode to Anode gap. Therefore the
OPTIMISED ELECTRON STREAM © value for the Suppressor Grid DC voltage will be the applied
Plate Voltage x 68% - but only up to a value 158% of that permitted by the
manufacturer's rated Screen-Grid Voltage (because in the case of Tube Type 803,
the maximum permissible Screen-Grid voltage is nominally 43% of the applied
Plate Voltage).

In other words, the
maximum applied DC Screen-Grid Voltage and/or Suppressor Grid Voltage for
OPTIMISED ELECTRON STREAM © operating
conditions will be determined by either the maximum rated Screen Grid or
Suppressor Grid voltage - whichever is the lesser.

IMPORTANT:

The SUPPRESSOR GRID (GRID #3) regulates
the flow of electrons in the tube in the same way as is the case for
Control-grid #1 and Screen-Grid #2.

In conventional "Pentode" configuration, the
Suppressor-grid is directly connected to the Cathode either internally by the
tube manufacturer or by the user.

Having left the Plate as surplus, randomly
travelling electrons, they find their way to the Suppressor-grid, thence
diverted to the Cathode to be absorbed back into the electron stream. This
arrangement obviously creates a short-circuit in respect of those electrons
attracted to the Suppressor-grid.

Thus there is an effective internal or external
circuit (as applicable) created between the Suppressor-grid and the Cathode,
that diverts some of the electrons back to the Cathode. This current is lost to
the output power and therefore reduces efficiency in the output power
stage.

In the case of the relationship between the
Screen-grid and Plate, most experts suggest that the Plate sees the Screen-grid
as the "Cathode", thus if this is so then the DC potential between Plate and
Suppressor Grid will be again determined by the linear distance between them -
unless the Suppressor-grid is purposefully connected to the Cathode.

It follows that in an OPTIMISED ELECTRON STREAM © technology amplifier
where the Screen-grid DC voltage is around 50% of the Plate voltage, if the
applied DC voltage is increased from 0 volts (Cathode potential) to a positive
voltage greater than the Screen-grid voltage, then the free electrons deflected
from the Plate will still be attracted to the Suppressor-grid because it is
still negative to the Plate.

It also follows that because the Suppressor-grid
is now positive to the Screen-grid, Plate current will increase - albeit
slightly.

Since both Screen-grid and Suppressor grids DC
voltages will be fixed, it becomes obvious that to control the electron flow
within permissible limits, the negative bias voltage applied to Grid #1 Control
Grid will need to be made MORE NEGATIVE.

It is also obvious that to limit the DC
current flow in Grid #3, and to prevent an AC signal short-circuit at the
Suppressor-Grids,  it is essential to load the Suppressor-grids by
installing a Grid-stopper resistor of around 75% of equivalent Plate to Plate
push-pull load impedance.

eg for a Plate to Plate load of 8,000 Ohms, the
transformer will present a load of 2,000 Ohms to each tube in the push-pull
pair.
2,000 Ohms x 75% is 1500 Ohms.
This is still a relatively small value so should not present significant voltage
drop or regulation issues.

This is an important difference between
convention and the OPTIMISED ELECTRON STREAM ©
configuration.

In a conventional pentode
circuit with grounded Cathodes, the Suppressor-grid is directly connected to the
Cathode so the Suppressor-grid is fixed at 0 VDC. However in the case of
OPTIMISED ELECTRON STREAM © configuration, the Suppressor-grid will be fixed at a relatively
high DC voltage set to enable the AC voltage to be aligned with the physical gap
in the tube. The circuit design then needs to accommodate this to protect the
Suppressor-grid from self-destruction through over-current.

The optimised AC load
impedance for the Suppressor-grid will be the ratio of DC Suppressor-grid volts
to Plate volts x Plate load impedance.

If the Suppressor-grid
load is provided by a fixed resistor the ratio for a Suppressor-grid - or set
of Suppressor-grids in one side of a push-pull set, will be the DC voltage ratio
x one quarter Plate to Plate load impedance

When used in an OPTIMISED ULTRA
LINEAR © configuration as shown above, it willalso will be loaded
by the tapped primary winding on the output transformer. The AC load impedance
will be determined by the ratio of the physical gap between the Cathode and
Suppressor-grid compared to the Cathode to Plate, as applicable - nominally 75%
of the Plate to Plate load.

The Suppressor-grid to
Suppressor-grid load impedance will be directly proportional, but the turns
ratio will be the square of the DC voltage ratio.

eg If the DC ratio is 75%
then the turns ratio for the Suppressor-grid tap on the output transformer will
be 0.75 x 0.75 = 0.5625 or 56.25%

Consequently,
proportionate audio POWER is drawn off from the Screen-grid and
Suppressor-grid by the output transformer.

PENTODE tubes suitable for high-fidelity
audio power output applications using OPTIMISED
ELECTRON STREAM © technology include:

802
803

804
837

I have not listed the EL34/6CA7, its smaller
brother the EL84/6BQ5 and cousin 6M5,  because the construction and effect
of the Suppressor-grid in these tubes is "nominal" and not in the same league as
those of the transmitting tubes listed above. Being manufactured from very fine
wire, the Suppressor Grid in these tubes is not capable of handling significant
current or power. For high-powered audio amplifiers
superior options are available as shown above.

However the
EL34/6CA7,  EL84/6BQ5 and 6M5 are suitable for effective use in
OPTIMISED ELECTRON STREAM © amplifiers
where the Screen-grid is set at 40% of Plate voltage and Suppressor-grid is
connected directly to the Cathode in the usual way, and in OPTIMISED ULTRA
LINEAR © amplifier designs.

3.1
OPTIMISED ELECTRON STREAM © TECHNOLOGY - SILICON DIODE FEED TO GRID
3

Consider also
OPTIMISED ELECTRON STREAM © TECHNOLOGY - SILICON
DIODE FEED (as described in Section 4 below) to
the Suppressor Grid.

In this case, a
conventional pentode configured amplifier - where the pentode tube has a
separate externally connected independent Grid 3 pin (typically connected to the
cathode or ground) - may be modified by installing a silicon diode between the
Suppressor-grid pin on the tube socket and Cathode or ground - as applicable.
The arrow must point towards ground - ie the marked terminal on the diode to the
Suppressor-grid pin.

This configuration allows
AC current to flow to ground but blocks DC current into the tube.

If any instability or
non-linearity occurs, a 1,000 Ohm resistor may be installed shunting the diode,
to create a permanent DC current path to the tube and anchor the Suppressor Grid
to the Cathode or ground.

4.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

SILICON DIODE FEED

Conventional historic tube
amp design using tetrode, pentode, or beam power tubes, evolve about a push-pull
output stage configuration typical of the sketch below:

In the example
cathode-bias is shown, however the same principles apply for a fixed-bias
design.

The notable feature of
this design is that everything is symmetrical and so is easy to draw, read and
understand.

The configuration of each
output tube is considered to be equally distributed about the central axis of
the output stage, (being the B+ and centre-tap of the output
transformer).

The core electronic design
concept assumes (deems) the Plate current in each tube will be identical at all
times and therefore any distortion in the plate circuits is cancelled out via
the push-pull effect. For an explanation of this refer to the RCA Tube
Manuals.

Screen-grids are not
considered to be more than a power output/efficiency enhancing benefit derived
from tetrodes v triodes, and as shown in the example above, are not connected in
any manner other than the most basic available - ie connected together and
supplied from the same power source as the Plates.

So in theory, all of the
components operate in a synchronous manner such that the output stage is
efficient and delivers its power at low distortion.

Unfortunately things are
not always as they appear.

The standard tolerance on
components is usually around + or - 10%, however some components, including the
tubes themselves, are assigned much wider tolerances.

Modern resistors are
typically + or - 5% however + or - 10% tolerance is still used.

Modern capacitors display
tolerances that vary with construction material and style.

Electrolytic capacitors
typically display -20% + 100% on nominal capacitance, thereby affecting gain
across the frequency range.

Vacuum tubes offer a
tolerance of + or - 20% on transconductance when new. Depending upon circuit
parameters, performance usually varies downwards with use. If one power tube
draws grid current early on then performance of the pair will
suffer.

Transformers vary
dramatically. Even if the primary centre-tap is exactly in the centre of
the total number of turns - ie the number of turns either side of the centre-tap
is exactly equal - the transformer may still not deliver an equal power
transformation between primary and secondary if the magnetic properties of the
two halves of the primary are not identical. Also, if the secondaries are not
equally distributed about the primary the induction into each secondary may vary
between halves. Factors such as core design and winding design will influence
the end result.

And what about the
symmetry of the driver stage? This too is subject to tolerance variables and
may deliver an asymmetrical signal drive voltage to the output tubes, resulting
in uneven power output from each half.

So what appears to be a
perfectly symmetrical design is in fact a widely varying practical
configuration.

One obvious variant is
Plate-current.

Many of the popular tube
types do not deliver a linear response between zero signal conditions and
maximum signal conditions. Whilst most guitar amp users would be familiar with
the concept of "re-biasing" after changing an output tube, it is not the norm in
hi-fi equipment. However despite good intent, guitar amp (and hi-fi amp) users
are blissfully unaware that their beautifully balanced output stage at zero
signal is nowhere near balanced at maximum signal.

In lower-cost/lower
quality transformers it will be found by measurement that the DC resistance of
one half of the primary is somewhat different to that of the other half -
resulting in uneven Plate voltage as Plate-current increases.

Imbalance in a push-pull
output stage results in loss of power and increased distortion.

This condition is more
likely in high-gain tubes like the EL34/6CA7, EL36/6CM5, KT88 and
6550.

SCREEN-GRIDS

Ideally, the output from
the Screen-grids is also identical and therefore cancels out at the output
transformer centre-tap.

But note from the above
design that any voltage appearing at the centre-tap of the output transformer -
whether it be derived from a difference in Plate characteristics, transformer
characteristics, untransformed signal, reflected back emf from the loudspeaker,
or simply hum as a ripple voltage - will appear on one or the other Screen-gid
as either direct injection or feedback.

Obviously Screen-grid
current will normally be a portion of (ideally balanced) Cathode-current, but
if different brand tubes - or tubes of differing design from the same
manufacturer - are used in the output stage pair it is likely that the
Screen-grids will behave differently - resulting in not only different tube
characteristics across the range but also differences in the Screen-grid
performances. Imbalance becomes apparent as distortion.

It follows then that a
useful object of amplifier design would be to eliminate the Screen-grid as a
variable from the system.

This proposition is very
well explained by Renato D. Tancinco of the Philippines, in his 1961 US
Patent 3153766

In this patent, Tancinco
presents his design, which aims to eliminate crossover distortion in tetrodes
and pentodes operating in Class B mode - EL34/6CA7 users take
note.

Unfortunately this design
only works in Class B mode because it cancels out part or whole of the opposite
polarity alternating signal in the output stage, however it does offer
significant advantage as the patent itself explains.

As noted in my
Screen-Grids paper linked above, reference to tube handbooks shows that in a
typical beam power tube, the Screen Grid current at maximum signal power is
around 20% of Plate current. This ratio of currents appears to be largely
independent of Plate voltage. It would therefore be reasonable to assume that up
to 20% of prospective signal power is lost in the Screen Grid circuit in a
conventional amplifier. (Power = volts x amps. Power supply voltage to the
screen-grids can be up to 100% of plate voltage)

(Note: Two notable
exceptions are the 807 and 814 beam power tubes that incorporate advanced
design technologies to increase tube efficiency and reduce distortion, however
in the overall scheme of things this technology appears to have been limited to
these two tube types - if you are aware of others please let me
know)

Two conventional options
present to overcome this nominal ratio of 20% plate Current:

The first is ultra-linear
connection, where all the electrons collected by the Screen Grids are fed into
the output transformer, but in the process modify the output stage
characteristics.

The second is to
increase the value of the Screen Grid-stopper resistor to a value
sufficiently high to resist the flow of electrons to the Screen
Grid.

However the Grid-stopper
resistor must be non-inductive to prevent oscillation. It must also be capable
of handling the Screen current passing through it. One way of doing this is to
use parallel carbon composition resistors (not film type - to prevent
fire) of sufficient number to obtain the required heat dissipation rating to do
the job without excessive temperature rise in the resistors.

A further problem here is
regulation of the Screen Grid supply. Obviously a Grid-stopper resistor of say
5,000 - 10,000 ohms will present a significant voltage drop when Screen Grid
current flows - if it does.

The loss of regulation may
be a price we have to pay to obtain a high standard of
performance.

Another and previously
unpublished option to creating an operating environment where the Screen Grids
will be at a DC potential sufficiently high enough to attract and accelerate
electrons towards the Plates but - to maximise power output - not to collect and
divert them to earth through the B+ supply, is the humble silicon diode
semi-conductor rectifier.

By inserting a standard
half-wave silicon rectifier diode in series with the Grid Stopper resistor, an
electronic control circuit is created whereby the Screen Grid will be able to be
energised at DC potential attracting and accelerating electrons towards the
Plate - still electrostatically controlling current flow in the normal way - but
blocking the flow of AC current from the Screen Grid back to the DC source - ie
"one way traffic"

This works because the
current flow in the tube is always from the Cathode to the Anode (Plate).
The diode, being a semi-conductor, blocks current flow in the reverse
direction, thus enabling DC current to feed it in the conventional manner but
blocks AC current from passing back through it to a load.

Thus then there is no
output circuit formed between the Screen-grid and the load so no current can
flow in the usual direction.

Don't believe it
?

See https://www.youtube.com/watch?v=CdADlclVMZM

A diode is a diode
!!

This causes the tube to
appear at first glance to behave like a triode. Note however that the tube is
still a tetrode, pentode or beam power tube as applicable - it is not a "super
triode" as some folks suggest. Plate-current is still controlled by Grid #1 and
Grid #2 voltages in the usual way - the difference being that the Screen Grid
signal current is not shorted to the B+ supply (AC ground) so is diverted to the
Plate and collected as significantly additional power output.

The diode is connected
between the B+ supply (line) and the Screen Grid (load) such that the arrow
points towards the Screen Grid. ie forward current is from the line to the
load.

This arrangement offers
huge benefits, because it prevents the Screen Grids from collecting electrons -
thereby diverting all the signal output to the Plates, increasing tube
efficiency, reducing distortion and increasing frequency response, as well as
eliminating the usual effects on changes in Screen Grid voltage on Plate Current
- therefore improving transient response.

One major benefit is
that the diode is not in the signal path and therefore does not modify the
sound.

A secondary benefit is
that there are no bypass or power supply capacitors (eg paper, polyester or
electrolytic) in the Screen-Grid signal path, which is a further major
improvement.

A further benefit is
that the non-linearity described under Fig. 2 above will be less of a problem
for us because the Screen-Grid component of the signal (ie those electrons
normally collected by the Screen-Grid)  is diverted to the
Plate.

Thus by inserting the
humble silicon semi-conductor diode in the output circuit, we can completely
break the bonds of traditional audio practice and take a great leap
forward!!

This is not tube heresy,
because the diode is not in the signal path - it merely prevents the signal from
being affected by adverse circuit parameters such as short-circuited
Screen-Grids, fluctuations in Screen-Grid voltage and power supply filter
capacitors.

The relatively low reverse
resistance of the diode appears to adequately satisfy the need for a low
impedance return path to AC earth, so the Screen Grid is not actually isolated
from AC earth - but there is sufficient impedance in the circuit to discourage
electron flow through it.

Note: Having regard to the
EIMAC articles regarding secondary emission in tetrodes referenced above, it
should be the case that the use of a silicon diode in the Screen-Grid supply
will not impede reverse current flow - provided a suitable bleeder
resistor is used between the diode and B+ source - as would be the case without
the diode in the circuit.

Some audiophile
experimenters have used single or series strings of zener diodes in this kind of
circuitry to regulate the DC Screen Grid voltage derived either directly
from the Plate or from the B+ supply, however I have used single conventional 1A
1000 PIV silicon rectifiers (from B+ supply only) with good
results.

Important Note: When
Zener Diodes are inserted in series with the Screen-Grid supply to both drop and
control Screen-Grid voltage, they are connected in reverse polarity to the
normal rectifier style diode described here, hence their effect on Screen-Grid
behaviour and of "sound" is quite different. A real danger with the series Zener
Diode configuration is that if the diode breaks down and short-circuits then
full supply voltage will be applied directly to the Screen-Grid. This may
destroy the tube in the process.

A further (and very
effective) enhancement is to use a separate diode for each Screen Grid (or each
set in parallel-push-pull) to ensure there is no cross-modulation in the
push-pull activity. This places back to back diodes between the Screen Grids,
which makes each half of the AC push-pull circuit independent to the other. Less
signal averaging and less cross-talk occurs between each half of the push-pull
pair, so the sound is cleaner.

In stereophonic amplifiers
using a common power supply, this system provides significantly greater channel
separation.

Of course, silicon diodes
can be retrofitted to an existing amplifier however the negative feedback loop
should be re-calibrated to suit the changed output circuit
conditions.

It may be necessary to
re-calibrate Grid #1 (Control Grid) bias to ensure Plate Current and Plate
Dissipation are optimised within the tube manufacturer's design centre
ratings.

In already set-up hi-fi amplifiers, it should
not be necessary to change operating conditions because the Plate current is
already determined by the Screen-grid voltage and not the Plate voltage - but a
prudent owner will check in any case to be sure.

If any reader can shed
further light on this breakthrough new concept please email your
thoughts.

Important: Please note
this modification is not suited to Class B guitar amplifier applications
(this means most of the "big" guitar amps) because the sound produced by the
silicon diode to Screen-Grids configuration is cleaner and less distorted,
dynamic range (transient response) is substantially improved and power output
is substantially increased - all advantages for hi-fi but not so good for a
guitar amp.

This is because many
guitarists, in order to attain or emulate a particular "sound" - and thus for
them "normal" usage of their guitar amplifier - operate the amplifier into the
severe distortion range by simply driving to full output or more. Some operate
in the sustained overload range continuously, using reverb, tape echo,
electronic echo or acoustic feedback as a musical effect.

In a typical commercial
guitar amplifier - particularly those with tube rectifiers - the power supply
will collapse and simply run out of puff when overloaded, resulting in
substantially reduced B+ supply voltage, lower power output and increased
(severe) distortion as the output signal goes into square-wave like response.
But the use of silicon diodes changes the tube characteristics insofar as the
normal limit on Plate current as controlled by the Screen-Grid is removed,
allowing Plate current to increase in proportion with the signal up to a maximum
"saturation" point where more drive in does not produce more power
out.

Note also that to maximise
output power and minimise distortion in a Class B amplifier, it is vitally
essential to balance each half of the push-pull pair of tubes to ensure
DC current in the output transformer is reasonably equal. The more tubes in
the output stage the harder this is to achieve.

Unfortunately, most
high-power (100 W RMS +) guitar amplifiers do not provide individual grid-bias
adjustments for the output tubes. Under such conditions, to achieve balanced DC
Plate-current it may be necessary to set Grid 1 bias control voltage at
maximum signal - not at quiescent (zero signal) as is popularly
expounded, and to mix and match the tubes either side of the output transformer
to attain reasonably equal balance in total Plate-current per side. (This may
result in some unbalanced DC hum at zero signal).

In an amplifier having a
single bias supply it may be also necessary to modify the circuit to install a
means for balancing AC signal drive voltage into the output tubes because, as
well as being dependent upon primary Grid #1 DC bias voltage, the Plate Current
will also be dependent upon AC input volts to Grid #1. Note though that the
downside to this is that when a tube is replaced, re-biasing is
essential.

For example, I conducted a
test with a Marshall Model 1959 100W super-lead amplifier, with 4 x EL34 tubes
running at 520 VDC B+.  When silicon diodes were fitted to the Screen-Grids
instead of the designed 1000 ohm grid-stopper resistor - one to each
Screen-Grid, power output increased from about 90W RMS in OEM configuration to
about 160 W RMS with diodes. Screen-grid current was negligible, suggesting the
tubes behave like triodes, however Plate current increased to about 180 mA per
tube, which increased net plate dissipation at full output to about 55 Watts - a
certain recipe for very short tube life considering the EL34 has a rated
Plate-dissipation of 25 W.

Plate current then in this
situation is primarily controlled by Grid #1 alone.

In this case the amplifier
simply gets louder and louder without the usual breakdown signs before clipping,
enabling the amplifier to be overdriven continuously to self-destruction. In
this particular amplifier, the power transformer is rated at about 250 VA, and a
quick calculation will show this component will also have a very short life
expectancy with silicon-diodes to the Screen-Grids - but the sound is
great!!

GENERIC CONCEPT
CIRCUIT

A conceptual circuit using the Type 803 Power
pentode Tube in an OPTIMISED ELECTRON STREAM ©
TECHNOLOGY design as described above, is
illustrated below:

This circuit is generic to all 3 Grid power
pentodes.

For Tetrodes and Beam Power Tubes
just delete the Grid #3 output transformer transformer tap, and its separate
Grid #3 DC supply - a Tetrode or Beam Power Tube will have only a Grid 2 in the
circuit.

Note: Westinghouse declared the following in
their June 1941 user instruction sheet supplied with each 803 tube:

Note the reference to "Tetrode" connection
and its effects.

This design feature offers a range of options
to the DIY designer/constructor.

Important Note: In the case of a Beam
Power Tube, the beam forming plates in a Beam Power Tube are not
normally connected to an active circuit element - ie are usually internally
connected within the tube to its own Cathode by the tube manufacturer. If
available as a separate connection, they should be externally connected to the
Cathode in the usual way.

Note: If a voltage measurement is taken either
side of the silicon diode in an ultra-linear configuration circuit - ie on the
screen-grid side and on the transformer side - obviously a reading will be
evident. The reading on one side will be out of phase with the reading on the
other side. DC current through the diode can be measured by inserting a small
resistor (10 ohms) in series with the diode on the supply side and reading the
current through the resistor.

5.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

PLATE, SCREEN-GRID (GRID
#2), SUPPRESSOR-GRID (GRID #3)
and
CATHODE AC CIRCUIT BYPASS CAPACITOR
.
.
5.1
OPTIMISED ELECTRON STREAM © TECHNOLOGY:

BYPASS CAPACITOR CONSTRUCTION AND
INSTALLATION

5.1.1    Non-polarised Bypass
Capacitor

Irrespective of choice of Screen-Grid and
Suppressor-Grid DC supply method, to obtain the full benefit from the
application of  OPTIMISED ELECTRON STREAM ©
TECHNOLOGY in Tetrode, Pentode and Beam Power Tube wide-band
amplifiers, it is absolutely essential to bypass (or wholly replace) the final
polarised electrolytic capacitor in ALL of the B+ Plate AND Screen Grid
AND Suppressor Grid supplies, with a suitable non-polarised high quality
mica, polyester, polypropylene or oil-filled paper etc. capacitor, having a
suitable value (of say 10 to 100 uF for audio), to provide an efficient and
stable AC bypass at all signal frequencies and under all operating
conditions.

The reason for the use of a non-polarised
capacitor is simply that polarised electrolytic capacitors offer asymmetrical
impedance to the current flow - depending upon the direction of the current
through the capacitor. Most importantly for high-fidelity reproduction,
polarised capacitors offer symmetrical characteristics to forward or reverse AC
current. This is evidenced by the widespread use of polarised capacitors in
loudspeaker crossover networks (which are purely AC) - even in the cheapest
commercial speaker systems.

The reason for the relatively large value of
capacitor suggested is that it is in series with the load on the Plate and
Grids, forming an LC or RC (depending upon the output stage configuration)
series network. If the value of C is low then peaks or resonance can occur in
the audio range - particularly in the mid to high frequency band where harmonics
are present.

This capacitor serves to effectively AC
short-circuit (or bypass) the DC power supply and thus eliminate the power
supply and its components from the AC signal path, and to ensure that any
shortcomings in polarised electrolytic capacitor performance are eliminated -
but in such a way that the signal is not significantly aurally affected.

In the case of the Plate circuit of Triode,
Tetrode, Pentode, Beam Power Tube and Ultra-linear wide-band amplifiers, the
non-polarised capacitor should be installed at the centre-tap of the output
transformer.

In the case of amplifiers having separate
Plate, Screen-grid and Suppressor-grid DC power supplies this requirement also
applies to each of the Screen-grid and Suppressor-grid DC circuits as
applicable.

To ensure adequate low frequency response
each of the separate circuits must have as large a value as is
practicable.

A 400 VAC (560 VDCW) Motor Run Capacitor is
ideal for this function and they are readily available in non-polarised
polypropylene construction. The voltage rating must be adequate to handle the
sum of the B+ DC voltage plus the AC rms signal output voltage.

I would recommend a parallel connected
non-polarised capacitor arrangement, to provide 100 to 200 uF total. For higher
B+ voltages the capacitors can be connected in series (800 VAC 1120 VDCW),
noting that when seriesed, the effective capacitance is halved.

A further refinement is to apply the "Rule of
Hundredth's", which says that instead of using a single capacitor - eg a large
electrolytic, a bank of capacitors is installed. Each capacitor is one hundredth
the value of its adjoining capacitor. For example, instead of installing one
single 100 uF electrolytic, instal a 100 uF, 1 uF, 0.01 uF, 0.0001 uF etc. wired
in parallel. Lead length of the smaller capacitors must be kept a short as
possible to prevent stray capacitance or inductance. It is also important to
ensure all capacitors installed are capable of coping with the applied voltages
without stress.

Of course the bypass capacitor is installed on
the line (source) side of the Screen-Grid-stopper resistor and/or diode, as the
case may be.

Note: This method may result in resonances in
the audio range, particularly to harmonics, resulting in sibilant accentuation
and "hissy" voice. It is thus not suitable in all amplifiers.

Note 1: Please note that the irrespective
of the value of the electrolytic filter caps - even to many thousands of uF -
the value of the non-polarised cap is still critical in relation to tone, or
spectral balance, across the audio range. To optimise tone experimentation is
essential.
Note 2: Modern "fast"
capacitors have a different tone to older oil-filled types. It may be necessary
to use oil-filled paper caps to obtain a smoother, less harsh tone. It all
depends upon the circuit design and componentry used. There is no definitive
answer. Unfortunately the older oil-filled paper caps are physically larger so
need more chassis space. They also do not usually come with flying leads, which
makes wiring more difficult because the terminal lugs are exposed. Ensure
voltage rating is adequate.

5.1.2    Electrolytic Bypass
Capacitors to B+ Supply

Please note that as explained in my Power
Supplies page, notwithstanding the obvious benefits from the use of
non-polarised capacitors as described above, it remains essential to good
transient response to instal large values of electrolytic capacitors to store
adequate power to satisfy the demands for transient peak signals.

The question is "how big a value"?

In the case of power requirements, full
particulars are provided in my Power
Supplies page.

But what about "sound"?

It has been demonstrated by early amplifier
designs produced in the 1940's and 1950's that the frequency response - even in
the highest quality equipment, tended to roll-off at both high and low
frequencies.

Frequency response usually deteriorated as power
output increased. This was an attribute ("power response") not usually presented
in the glossy sales brochures.

Modern digital recording techniques and playback
media have dramatically increased both low and high-frequency response for the
typical recording, challenging even the very best of tube audio amplifier
designs - particularly when played at high-volume.

But all is not lost!!

Modern capacitors offer improved performance and
reliability over their ancestors. They are also available in values unheard of
in the days of tube rectifiers. (eg 32 uF was "large" in the 1950's but now
100,000 uF is "medium.")

The thin film caps of today charge and discharge
much faster, enabling larger values to be used in common applications where
large values were previously forbidden - such as interstage coupling
caps.

Modern capacitors offer reduced unwanted side
effects, such as inductance, leakage and resonance, so offer improved
high-frequency performance and audio clarity.

One effect of this is a reduction in the level
of negative loop feedback needed to offset roll-off in frequency response of an
amplifier - a definite advantage.

In the case of low frequency performance we can
retrospectively improve old designs by using higher values of filter bypass
capacitors.

Since the bypass capacitor forms the return AC
circuit for each stage in the amplifier, it follows that the impedance of the
bypass (filter) capacitor at any given frequency will be a portion of the total
impedance of the circuit being bypassed. Now since the impedance of a capacitor
varies in direct proportion with the value of capacitance at any given
frequency, it follows that providing we reduce the value of the capacitor's
impedance to a value that has minimal effect upon the circuit, then we can
attain improved performance from the circuit.

Another way of explaining the concept is that
traditional tube electronic design engineering principles assume that the B+
rail is at AZ zero (or earth) potential - regardless of the applied DC
voltage.

This is a practical approach for many
applications, but ignores the reality that capacitors - particularly those of
the electrolytic variety - have their own characteristics, which are injected
into the circuit and therefore MUST influence the sound we hear.

Since the capacitive reactance of a 25 uF
capacitor at 30 Hz is nominally only 212 Ohms, it suggests that the influence of
the capacitor may be negligible. However if the inductance of the capacitor is
only 0.1H then the inductive reactance in the capacitor is 3140 Ohms at 5 kHz -
and more at higher frequencies.

So we can see that increasing capacitive
reactance will decrease low frequency performance and decreasing capacitor
induced inductive reactance will increase high-frequency performance.

Since measured inductance and capacitance values
from typical large capacitors - eg above 500 uF - do not appear to display the
characteristics experienced aurally there may be an alternative explanation. It
may be that the bypass capacitor forms an RC or LC series network with the plate
resistor and/or output transformer, forming a bass boost circuit. This is easier
to see with the output transformer, where either one end (in the case of a
single-ended output stage) or the centre-tap (in a push-pull output stage) forms
the terminal at which the output is taken off. but in the case of driver stage
it would seem the series decoupling resistor causes the formation of the RC
network that creates the bass boost effect.

Either way it is desirable to tune the network
to the preferred frequency - eg 40 Hz

The question is what is the magic number?? How
big to we need to go??

To attain extremely good low-frequency response,
from practical experimentation I would suggest the minimum value of the
bypass capacitor for any stage in an amplifier or pre-amplifier, including the
output power stage, may be calculated by dividing the constant 15,000,000 (15
million) by the value of the plate load resistor (or cathode-follower cathode
load resistor) for that stage.

This approach will produce values of 150 uF for
a 100k plate resistor and 2,250 uF for a 6,600 Ohms Plate to Plate output
stage.

If those values appear frightening, then try a
lesser constant - say 10,000,000

If the B+ supply supplies more than one stage
or, in the case of some phase-splitters having more than one plate resistor,
then the value of the resistor should be calculated as the average of all the
resistors in the circuit.

If the B+ supply supplies a stereo pair of
amplifiers then the value of the plate load resistor used to calculate should be
half of the single channel value - ie each bypass capacitor should be twice the
size as for a single channel.

If the output stage uses more than one pair of
tubes from a single transformer, then use the transformer actual nominal rated
load impedance - not the effective impedance as seen by each pair.

It will be noticed that these values of
capacitance are substantially higher than convention, however this is what I
have determined form extensive critical listening tests.

The general object of this design approach is to
introduce a circuit resonance induced tonal characteristic that is
pleasing.

Too high a value of capacitor will deliver
undefined bass, so some tweaking may be necessary.

It is desirable for all stages to have the same
tonal characteristic, so this formula assists to achieve that. In other words,
each stage should ideally have proportionately the same tonal or frequency
characteristic.

Note 1:  This "rule of thumb" formula deals
only with tonal characteristics - power (energy) requirements for the output
stage are not covered by this. However it is easily seen that the values
produced by this design approach will supply adequate power for most
applications.

Note 2: Regardless of the value of electrolytic
capacitors used for bypassing and filtering, non-polarised bypass capacitors
are still essential for good high-frequency performance and low intermodulation
distortion.

Note 3: In the case of guitar amplifiers the
above formula is not applicable because the lowest frequency to be reproduced is
about 80 Hz. In this case a constant of say 10,000,000 or less would be
appropriate.

Note 4: In the case of bass guitar amplifiers,
where the lowest frequency to be reproduced is around 40 Hz, then for
outstanding results the constant needs to be in the region of 20,000,000. This
value of capacitance will also deliver adequate power to the output
stage.

Note 4: Large values of capacitor can be lethal,
so discharge resistors should be installed as per instructions provided in my Power
Supplies page.

Note 5: In my experience, larger bypass
capacitors have the effect of improving circuit stability and improving
high-frequency audio performance.

5.2     OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

RATIO OF SCREEN-GRID AND SUPPRESSOR-GRID BYPASS TO PLATE BYPASS
CAPACITORS

5.2.1     Introduction

Since the advent of
Tetrodes and Pentodes, it has been standard practice to instal a small bypass
capacitor from the Screen-grid to ground (Cathode) in Tetrodes and Pentodes used
in voltage amplifier stages.

This is shown as C2 in the
figure below.

The purpose of the bypass
capacitor is to reduce Grid to Plate capacitance, remove undesirable audio and
high-frequency signals such as RF components from the output before the load,
and to improve the decoupling and stability of the stage.

In voltage amplifier
(driver) stages the value of this capacitor has historically been in the order
of 0.1 to 0.5 uF, the latter value being considered by Radio Engineers to be
adequate for good quality audio purposes. A similar situation exists in RF power
amplifier stages.

It should be noted though,
that this value of capacitor is usually associated with small tubes in
high-impedance circuits, such as the EF86, 6AU6, 6U8, 7199 etc, where the value
of Screen-grid supply resistor may be in the range 100k to 1 Meg Ohms with very
low Screen-grid current, .

An example is shown in the
Mullard
High-fidelity Pre-amplifier.

5.2.2
Theory

For the theoretically
minded, the formula
for calculating the value of the bypass capacitor in a voltage amplifier stage
is given courtesy of the Radiotron Designers Handbook, 3 rd Edition
(1940).

5.2.3
High-fidelity Audio POWER AMPLIFIER Applications

Notwithstanding the above
historic convention, in high-fidelity audio POWER AMPLIFIER output stages
a very different situation applies.

If we think for a moment,
it can be easily seen that there is both a DC and an AC signal path from the
negative terminal of the bypass capacitor up through the capacitor to the
Screen-grid external to the tube, thence from the Screen-grid to the
Plate inside the tube. The latter will be the case (even if we do not
want it) because the Screen-grid is negative to the Plate.

As explained above, the
portion of Screen-grid current in a power tube can be quite high - depending
upon output stage configuration and applied voltages to the Screen-grid and
Plate respectively.

Since the internal
impedance of the bypass capacitor will be relatively small and the Screen-grid
is operating independently of Control Grid #1, it follows that the magnitude of
the current flowing from Screen-grid to Plate will depend more or less entirely
upon the actual applied DC voltage between Screen-grid and Plate and the value
of the Plate load - which will be also seen by this secondary
circuit.

That is "secondary" to the
primary Cathode to Plate circuit.

Obviously, the higher the
difference between actual applied DC voltage between Screen-grid and Plate, the
more current will flow.

In Tetrode and Pentode and
Beam Power Tube applications where the Plate and Screen-grid operate at the same
DC voltage, including Triode and Ultra-linear connections, in the conventional
and very common configuration shown above, it is suggested by most writers that
the Plate will function as the primary anode so long as the Plate signal voltage
does not drop below the DC Screen-grid voltage. In this case, tube manuals show
that about 10-20% of Cathode current is lost in the Screen-grid
circuit.

Since the bypass capacitor
C1 is common to both Plate and screen-grid circuits, in terms of frequency
response and dynamic response whatever happens in one will happen in the
other.

But when the DC
Screen-grid voltage is less than the Plate voltage - as in RF linear amplifiers,
transmitter modulator amplifiers, transmitting tube audio amplifiers, Public
Address amplifiers or OPTIMISED ELECTRON STREAM
© TECHNOLOGY applications as shown in the
following diagram - then current MUST flow between Screen-grid and
Plate.

This is because a
secondary DC and AC circuit is established between Screen-grid and Plate, with
the Screen-grid forming the negative terminal/element.

In high-voltage amplifier
designs, the voltage between Screen-grid and Plate may be in the order of
several kilovolts (kV).

Since the lower leg from
Cathode to Screen-grid is through the bypass capacitor, and that is external to
the tube, it follows that the limiting factors to current flow in this secondary
circuit will be the Plate load impedance for AC current (the DC resistance of
the output transformer is negligible) and the value of the Screen-grid resistor
if used (either voltage dropping resistor or grid-stopper) for both AC and DC
current.

For explanatory purposes,
the internal tube resistance between Screen-grid and Plate may be assumed to be
zero.

It follows then that the
Cathode to Screen to Plate secondary AC circuit is in parallel with the Cathode
to Plate primary AC circuit.

As explained above, the
value of Screen-grid current - and therefore its contribution to audio power
output, can be significant.

Since the current in both
circuits combine together in the negative to positive Screen-grid to Plate
section of the circuit inside the tube (and thence common return to the Cathode
via the B+ filter capacitor at the output transformer centre-tap) it follows
that any difference in the audio signal between
primary and secondary circuits will be apparent at the output
transformer.

For example, if the
Screen-grid bypass capacitor is too small, that portion of audio power output
contributed to by the Screen-grid secondary circuit as described herein, will
not have the same low-frequency response as the primary Plate circuit and
therefore low-frequency power output will be proportionately
reduced.

For example, in the Mullard
High-fidelity Pre-amplifier circuit, typical of conventional design, the
Screen-grid bypass capacitors C9 is 80 times the value of C8, and C17 is 160
times the value of C12.

The value of the
Screen-grid bypass capacitor will also affect the operation of the Plate
circuit B+ bypass capacitor, because the B+ Screen bypass cap and B+ Plate
bypass cap are in series in the signal circuit.

Hence it may be deduced
that:

Where a separate
Screen-grid power supply is provided,  it is most important for full-power
hi-fi reproduction at very low frequencies, and for signal balance within the
tube, to ensure the value of the final Screen-grid B+ bypass filter cap is not
less than the value of the Plate circuit B+ bypass filter cap.

In the above diagram the
Plate bypass capacitor is shown as C1 and the Screen-grid bypass Capacitor as
C2.

Note: Even though the
installation of a voltage dropping resistor or grid-stopper resistor to the
Screen-grid (R3 in the voltage amplifier circuit diagram above) may reduce AC
and DC current, the issue of AC impedance remains - hence the grid resistor may
be disregarded for this aspect of hi-fi design.

It will be readily seen
that in a power output stage where the Plate and Screen-grid share a common AC
circuit return at the power supply, and regardless of the use of a grid-stopper
or dropping resistor (R3 in the above diagram) or not, the above argument does
not apply because both Plate and Screen-grid will share a common B+ bypass
capacitor.

BUT - where the
screen-grid is supplied through a filter choke from the centre-tap of the output
transformer at C1, and bypassed by its own electrolytic capacitor C2, the
configuration shown above is still is applicable.

However bear in mind that
the regulation of Screen-grid voltage is also extremely important so a separate
supply should remain an essential design element.

The above comments also
apply to the Suppressor-grid

5.2.4    Cathode Bypass
Capacitor

In the case of Tetrode or Pentode amplifier
output stages having Cathode Bias, it is usual to instal a Cathode-bypass
Capacitor as shown in the following diagramme:

.
It
can be readily seen from the way the circuit is drawn that the Cathode-bypass
Capacitor is in series with the B+ Plate and Screen-grid supply.

It follows that if the
Cathode-bypass capacitor is too small, that portion of audio frequency response
determined by the Cathode circuit will not have the same low-frequency response
as the primary Plate circuit and therefore low-frequency power output will be
proportionately reduced.

From the above explanations it can be
demonstrated that the value of C3 must be at least equal
to the value of C1.

The more theoretically minded can calculate the
actual values needed for equilibrium in the frequency response characteristics
for each part of the circuit.

Where a separate Screen-grid power supply is
used, as shown below,

it can be readily seen that the Plate and
Screen-grid bypass capacitors C1 and C2 are in AC parallel but the set of both
is in series with the cathode bypass capacitor C3.

Hence it is also essential that the nominal
capacitance value of C3 is equal to the SUM of C1 + C2.

Where separate Cathode-bypass capacitors are
used to each Cathode, the above rule still applies for each capacitor.

It is usual for Class A amplifiers to use a
single common cathode resistor and no bypass capacitor per push-pull pair of
tubes - this is obviously more AC linear.

5.2.5 Power Factor

Ideal values for C1 and
C2 and C3 can be in the region of up to 5,000 uF per push-pull pair of  tubes -
see warning re risk of electrocution.

To maintain
OPTIMISED ELECTRON STREAM © TECHNOLOGY
design standards in the output stage circuit,
some circuit tuning is essential to optimise the "POWER FACTOR" in the output
stage circuit.

In terms of "sound", this
is a very critical issue.

This extra step is needed
to tune resonances in the output stage circuit to match the characteristics of
the output tubes, output transformer and loudspeakers to the signal "sound" or
"tone" and listening room acoustics. Too low a resonance may result in a "dull"
or "flat" bass sound. Too high a resonance may result in "bass boom" and loss of
definition.

To optimise the output
stage circuit and deliver maximum power from minimum output impedance with
maximum loudspeaker damping characteristics, changes in power factor
produced by the inductive reactance in the output transformer will ideally be
cancelled out by the capacitive reactance in C1.

To optimise the value of
C1 some degree of practical "trial and error" experimentation is needed - a
frustrating experience but one worth the effort.

Note C2 and C3 need to be
suitably modified as described above and hereunder to maintain the correct
ratio to C1 - ie the same capacitance value.

Note also that to overcome
non-linearity in the AC bypass circuit, C1 and C2 may need to be bypassed by a
small stabilising capacitor in the region of 0.5 uF - but since this small
capacitor will directly affect the high-frequency response, it too must be
chosen with care regarding both size and material of construction.

IMPORTANT:

All of the Plate
Circuit output stage power passes through the AC circuit formed by the output
transformer and C1 - and whatever else shunts C1.

Since L1 is directly
connected to the Power Supply, it follows that the Power Supply is in series
with L1 and shunts C1.

This is illustrated in
the diagram below:

So to minimise the
effects of the Power Supply rectification and filter circuitry on "sound", the
value of L1 should be as large as can be practicably sustained - noting that the
DC resistance of L1 will directly reduce the available B+ voltage to the
Plates.

The larger L1 is, the
more AC current will be forced through C1 and less through L1 and associated
circuitry.

Another way of
expressing this is to say that whatever AC power is lost into L1 and its
associated source bypass components - all of which shunt C1 - the less linear
the output stage will be at low frequencies.

An inductance of about
10 Henries (minimum) is desirable for L1.

This also holds true
for simple one stage filter choke systems, because the rectifier/filter is
always shunting C1 - thus will always affect its performance and effect on
"sound".

IMPORTANT: To prevent
instability in the power supply (which dramatically affects audio sound as heard
through the loudspeaker) it is desirable to ensure C2/L1 and C3/L2 are of equal
value. Where only C2/L1 are used then it is desirable that C1 and C2 are of
equal value. It is thus obvious that a three stage filter enables a higher value
of C1 than for a two stage filter.

This point will be
expanded in future revisions of this page.

5.2.6
Silicon Diode Feed

When silicon diodes are
installed in the Screen-grid and Suppressor-grid circuit when using my
OPTIMISED ELECTRON STREAM ©
TECHNOLOGY design, the above bypass
requirements still apply because although the diode will block AC from passing
from the Cathode to the Screen-grid and Suppressor-grid internally, it cannot
stop the current flow in the external secondary circuit described
herein.

5.2.7
Voltage Doubler Power Supply

Where the Screen-grid B+
supply is taken from the output of a voltage doubler rectifier, it may be the
case that for convenience the Screen-grid supply is taken from the mid-point of
the two series-connected filter capacitors.

However if we analyse the
effect of this arrangement having regard to the above, it is easy to see that
the external Screen-grid bypass current is passing from negative to positive
through the lower filter capacitor.

It is also easy to see
that return external Screen-grid bypass current is passing from positive to
negative through the lower filter capacitor.

Consequently, it is
evident that the two currents will cancel out.

Bad move!!

So in other words, this
configuration is not desirable for high-fidelity reproduction, even though it
may appear to work satisfactorily in public address amplifiers.

There is also an issue of
the effect of differences in capacitor characteristics in the forward and
reverse directions, as well as the effects of distortion caused by the output
transformer.

There is also the issue of
excessive ripple in the DC supply.

One solution to both
issues may be to instal a 60 mA filter choke in the line (supply) side of the
Screen-grid B+, then add a second filter capacitor to the Screen-grid B+ at the
load side, to provide a separate path for forward and reverse
currents.

The value of the second
(final) bypass capacitor should be equal to or greater than the value of the
seriesed pair.

5.2.8
Capacitive Voltage Dividers

Where the Screen-grid B+
supply is taken from the centre-point of a pair of series-connected electrolytic
capacitors in the Plate B+ circuit, the same problems arise as explained
above.

NOTE: To ensure equal
voltage distribution over time (service life) series connected electrolytics
MUST be stabilised for equal voltage distribution by means of an external
circuit system such as shunt resistors (voltage divider network) or power
supply configuration.

5.3     OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

SCREEN-GRID AND SUPPRESSOR-GRID BYPASS CAPACITOR SIZE:

In the case of
OPTIMISED ELECTRON STREAM © TECHNOLOGY where a separate Screen-grid and Suppressor-grid power supply is
provided, for full-power hi-fi reproduction at very low frequencies it is
essential to ensure the value of the final Screen-grid B+ bypass filter cap (and
Suppressor-grid in the case of a Pentode) is NOT LESS THAN the total effective
value of the the sum of all Plate circuit B+ bypass filter capacitors directly
connected to the centre-tap of the output transformer - polarised and/or
non-polarised.

Where Cathode-bias is
employed, in addition to the above it is also essential to ensure the value of
the Cathode-bypass capacitor(s) is not less than the value of the Plate and
screen-grid supply capacitors

Note: Where one or more
filter chokes are used in the B+ supply, the value of capacitors before the
choke(s) may be ignored for this requirement.

5.4
NEGATIVE LOOP FEEDBACK

Where negative loop feedback is used from the
loudspeaker terminals back to an early voltage amplifying stage, it can be seen
from the above that any defects in performance in the output stage will simply
be transferred back to the input, with the result that the amplifier will always
be in a constant state of correcting itself.

In real life we have the following variables in
the power stage system:

output tubes having unequal and non-linear
characteristics over their minimum to maximum Cathode current range - ie
Plate current v Control Grid voltage

the Control Grid "grid current" varying with
AC signal amplitude,

the Control Grid AC bias voltage varying with
AC signal amplitude,

in ultra-linear systems the Screen Grid
voltage varying with AC signal fed back from the output
transformer,

the Screen Grid current varying with the AC
circuit voltage between Screen and Plate,

the Plate, Screen and bias DC voltages
varying with total load on the common power transformer - ie transformer/power
supply "regulation".

unequal characteristics in each half of the
output transformer,

output transformer frequency power
response is less than the full audio range available at the input
terminals,

back EMF from the loudspeaker reflected
(transformed) into the Plate circuit (loudspeakers have different "push" and
"pull" cone characteristics due to suspension construction and enclosure
characteristics) (Note the cone is tapered in the "pull" direction and
assisted by air pressure however it is cup shaped in the "push" direction) -
ie is acoustically non-linear. (This comment does not apply to planar
loudspeakers)

non-linear impedance loading by the
loudspeaker over the full frequency range - typical 8 ohm speakers reflect
frequency dependent loads between 6 ohms and 100 ohms - ie reflected Plate
circuit load can increase from the nominal by a factor of 10 to 12
times

non-linear negative feedback resulting from
non-linear audio frequency output volatge (compared to the input
terminals)

reflected plate load varying with
frequency

loudspeakers present reactive loads to
the amplifier - this contrasts poorly with laboratory engineering performance
tests using fixed resistive loads

efficiency losses (insertion loss) in the
output transformer - at least 1 db average but not necessarily linear across
useable audio frequencies

.
Hence any use of negative loop feedback should be
deferred until the output stage is optimised according to the requirements of my
OPTIMISED ELECTRON STREAM ©
TECHNOLOGY

The best plan is to forget it and live with the best option without it.

Please refer to my
SCREEN-GRIDS page for full particulars of this requirement.

6.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

CONTROL GRID (GRID #1) AC CIRCUIT BYPASS
CAPACITOR

In output stages of audio amplifiers it is
common to use FIXED BIAS or BACK BIAS to apply and control the
Grid #1 - Control Grid bias voltage in the output tubes.

Designers always consider the requirements for
direct current operating conditions but often ignore requirements for
alternating current conditions - ie signal voltage.

The Grid #1 Resistor to each output power tube
forms part of the load for the preceding driver stage. Hence for maximum
efficiency and stability, the return path from the output of the driver stage
circuit back to its Cathodes should be direct and have very low
impedance.

6.1
OPTIMISED ELECTRON STREAM © TECHNOLOGY Principles

Bias, by definition, requires a voltage
potential to be present between the Control Grid and the Cathode.

CATHODE BIAS usually results in the Control Grid
being at nominally 0 VDC and the Cathode at the required bias voltage being +
VDC.

FIXED BIAS usually results in the Control Grid
being set at the required bias voltage being - VDC and the Cathode at nominally
0 VDC. In the case of FIXED BIAS it is essential to bridge the difference
between the central axis of the AC signal input and the power tube Cathodes,
such that the central axis is at the same AC voltage as the power tube
Cathodes.

It is essential for stable high fidelity
operation to ensure that:

a)    the Grid #1 circuit has a
reliable and predictable low-impedance return path for the AC signal
voltage from the preceding stage
b)    the Grid #1 circuit has a reliable
and predictable low-resistance return path for the DC bias voltage
between Grid #1 and the Cathode of the same tube.
c)
in a balanced driver stage for Classes AB, AB1, AB2 and B, where the power tubes
are biased towards cut-off and driver stage output signal voltages are
symmetrically balanced about a virtual central axis having a nominal potential
of 0 VAC, it is imperative to ensure the junction of the two Control Grid
resistors of the output tubes be AC grounded to 0 VAC.

Thus in an RC
(resistor/capacitor) coupled amplifier, it is a fundamental requirement that
the Grid #1 Resistor to each power tube provides a return circuit path to
the Cathode of  BOTH the preceding driver tube AND the power tube to which
it is connected.

This is usually, but not always, through the
earth or ground terminal of both AC and DC applied circuits.

Fortunately, in most amplifiers using FIXED BIAS
or BACK BIAS, the Cathodes of the output tubes are directly earthed (to the
chassis), thereby providing a convenient return circuit at 0 VAC
potential.

6.2    Cathode Bias

In most amplifiers using CATHODE BIAS, the
Cathodes of the driver tubes are earthed either through a bypassed or unbypassed
Cathode resistor, hence the Cathode terminal is indirectly AC earthed to
complete the return circuit for the driver tube(s).

This principle applies particularly to CATHODE
BIASING of power tubes, where the Cathode terminals may well be at say
+50 VDC but simultaneously at  0 VAC. . (An exception is where the output
stage is in Class A and the common cathode resistor is unbypassed to
develop negative current feedback in the Cathode circuit.)

Typical Ultra-linear output stage configuration
with CATHODE BIAS. Note the bypass capacitor on the common Cathode resistor,
thereby providing a low-impedance AC path to ground.

Typical Triode-connected Pentode output stage
configuration with adjustable CATHODE BIAS.
Note
the absence of a bypass capacitor on the common Cathode resistor.

This circuit features + or - adjustment of
common bias voltage and also fine adjustment of balance between the
tubes.

6.3    Fixed Bias
Adjustment

It is common practice in FIXED BIAS amplifiers
to incorporate an adjustable bias control circuit, incorporating an adjustable
resistive network to enable precision adjustment to the Control Grid voltage
and/or Plate Current of the output power tubes.

Typically, this negative polarity DC voltage is
sourced from a half-wave or full-wave tube or solid-state rectifier, filtered by
a simple resistor/capacitor network.

Typical Fixed Bias amplifier incorporating
adjustable bias to the power tubes.

Note: In this design, the 20 MF (uF) capacitor
serves to bypass the AC return circuit from the driver stage to ground, thereby
providing a direct return circuit path to the Cathodes of the driver tube. This
capacitor is connected in such a way as to ensure the Control Grids are equally
earthed at all times through their respective Grid Resistors, regardless of
applied DC bias voltage. It also ensures the central axis of the balanced signal
input is at the same potential as the Cathodes - an extremely important
component.

6.4    Bypass Capacitor
Material

In commercial amplifiers with FIXED BIAS, to
minimise cost the bypass capacitor is nearly always of the polarised
electrolytic variety, but this means that the AC return circuit is not
symmetrical. (Electrolytic capacitors have different characteristics in
positive and negative polarity circuits).

Consequently, to ensure
symmetrical AC circuit configuration it is essential to instal a suitable
non-polarised bypass capacitor into the circuit at a point closest to the Grid
#1 of the output tubes as is practicable.

Typically this will be at the junction of the
two (or more) Grid Resistors.

If a driver transformer is used, and its
centre-tapped secondary is not directly connected to the power tube Cathode
circuit, then instal the bypass capacitor between the centre tap of the
secondary and earth.

It is therefore absolutely essential to bypass
(or wholly replace) the final polarised electrolytic capacitor, with a suitable
non-polarised high quality mica, polyester, polypropylene, paper or oil-filled
paper capacitor, having a suitable value (of say 1.0 uF or a 10 uF or more motor
start capacitor as above for audio), to provide an AC bypass at all signal
frequencies and under all operating conditions.

This small capacitor serves to effectively
short-circuit (shunt or bypass) the DC power supply and thus eliminate the
power supply and its components from the AC signal path of both the driver
tube and power output tubes, to ensure that any shortcomings in polarised
electrolytic capacitor performance are compensated - but in such a way that the
signal is not significantly aurally affected.

Importantly, it also provides an automatic
safeguard against the adverse effects of poorly contacting bias potentiometers
and/or adjustable wire-wound resistors.

Normally, the value of the bias supply capacitor
will not exceed 10 uF (to ensure fast charging to full bias voltage before the
output tubes heat up and commence to conduct Cathode Current) so an extra one or
two uF will not significantly affect the charging circuit performance.

If preferred, to achieve the same end result
as described above, the final Control-grid power supply filter polarised
electrolytic capacitor (and the first as well if so inclined) can be wholly
replaced with a polyester or paper or motor start capacitor of say 8 uF value
and having a suitable DC voltage rating.

Suitable capacitors may also be reclaimed from
unwanted fluorescent lamp-holder assemblies.

For the ultra-fastidious, the "Rule of
Hundredth's" may also be applied.

Note: There is no practical limit as to the
value of the bypass capacitor, provided the bias supply is capable of charging
it quickly to ensure bias voltage is present when the power tubes warm up and
commence conducting. I have successfully used values around 100,000 uF, shunted
by suitable non-polarised polypropylene caps.

7.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

SINGLE OR DOUBLE STAGE PI FILTER
IN EACH POWER SUPPLY
FILTER
CAPACITORS OF EQUAL CAPACITANCE
EACH
INDUCTOR FILTER CHOKE BYPASSED BY A REVERSE BIASED SHUNT DIODE (SNUBBER
SYSTEM)

The concept is shown
here:

.
The purpose of this configuration is to stabilise the voltage
into the B+ rail. (Note: "Stabilise" means free from
self-oscillation)

Ripple voltage appears at
the input.

The ripple voltage
includes harmonics from the mains and spikes from the rectifiers - both of
which pass through the inductor filter choke and appear at the output
transformer centre-tap (AC earth) - so thereby appear in the output stage
circuit as a signal modulating voltage and, by transformation, audible in the
loudspeaker as distortion.

The shunt diode
short-circuits the choke in the reverse direction

Any unwanted spurious
voltage - such as switching spikes - are either eliminated from the system or
dramatically attenuated

The use of a small
capacitor or capacitors connected in parallel with the first electrolytic cap
provides a path for spikes and any other mains harmonics etc to be diverted to
the negative rail.

The output voltage is
therefore, for practical purposes, spike free

The capacitors should be
of equal value and type so that the two capacitors operate in sync with the
input voltage - ie the current in is matched by an equivalent current out,
synchronised in time. That is to say the charge and discharge rates over time
should be equal. If they are not equal, higher than normal rectified ripple
mains frequency (ie 100 Hz or 120 Hz) ripple spikes can be generated within the
filter system.

Capacitance values up to
20,000 uF work well for me.

NOTE: See warnings re
risk of FATAL electric shock from charged capacitors.

The benefit of the PI
filter is that even a small value of choke inductance does wonders for
substantially reducing ripple.

Elimination of Ripple is
vitally important for true hi-fi because the B+ rail is the AC ground for the
signal.

If this voltage "dances"
around with spurious ripple signals and/or the actual centre-point of the
output transformer not being at its nominal zero AC volts true centre, then the
output will be modulated by this voltage. It can either be bypassed to ground
via the filter capacitors or transformed to the loudspeaker (in which case it
will appear in the negative feedback system - if used)

This system is very stable
and enhances clarity and frequency response

8.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

DECOUPLING OF THE POWER
AMPLIFIER FROM THE POWER SUPPLY
BY MEANS OF
A SERIES CONNECTED SILICON DIODE

The conventional B+ supply looks something like
the following diagram:

It is explained above why Filter Capacitor C1
forms a vital part of the AC circuit in the output stage.

It is also explained why it is desirable to
"optimise" the value of C1.

Further, it is explained that the Power Supply
is connected in parallel with C1 and therefore also forms part of the AC circuit
of the output stage.

Consequently, it can be easily demonstrated that
the "sound" of the amplifier will be directly affected - maybe adversely - by
the characteristics of the Power Supply.

Since the object of OPTIMISED ELECTRON STREAM © TECHNOLOGY
is to "optimise" both the performance and sound
of the amplifier, it follows that if the Power Supply can be isolated, or
decoupled, from the power stage then a positive benefit might
result.

This object may be
easily and economically realised by installing a series connected silicon
diode rectifier into the B+ supply - having its arrow pointing towards the load
- between the Power Supply output terminals and C1 - ie
AFTER  the Power Supply and BEFORE C1.

Note that C1 must
always remain part of the Amplifier, because it forms a direct path in the AC
output circuit.

So regardless of whether
the Power Supply is of the simple centre-tapped full-wave or full-wave bridge
rectifier or full-wave voltage doubler rectifier, or it has a capacitor filter,
choke filter or some other combination of filter - or even includes a voltage
stabiliser or regulator - to ensure the OPTIMISED VALUE OF C1 is not altered by
shunting the Power Supply circuit, significant improvement to performance can be
made by installing a series silicon rectifier diode as described.

To ensure cool operation
and reliability, the current rating of the rectifier should be generous in
relation to the power requirements of the amplifier - eg 3A or 6A. The rectifier
can be of the same specification as that used in the Power Supply
B+.

Where a separate
Screen-grid Power Supply is used, then the same principles apply to
C2.

Where an inductor - such as L1 above - is used
in the Power Supply, it is preferable for the diode to be installed after the
inductor - to ensure the Power Supply is completely isolated from the
amplifier.

This method makes it easier to adjust the tonal
balance or "tone" of the amplifier power stage - because fewer components are
now in the signal path - therefore there are fewer interactive variables
to deal with when changing component values.

The same principle apply to L1 and C2.

It will be seen for the above that in the case
of simple filters having just one capacitor after the rectifier - and nothing
else - the requirement described is met. Unfortunately that simple configuration
results in significant hum and ripple and poor quality direct current (DC) - so
is not recommended for high-fidelity amplifiers.

9.    OPTIMISED ELECTRON STREAM ©
TECHNOLOGY:

DECOUPLING OF
EACH STAGE OF THE VOLTAGE/POWER AMPLIFIER
FROM ITS NEXT-FOLLOWING ADJOINING
STAGE BY MEANS OF A
SERIES CONNECTED SILICON
DIODE

Having described how decoupling of the amplifier
from its power source facilitates and enhances performance, it follows that the
same principles apply to driver stages.

In a conventional amplifier, the stages are
connected in a "cascade" configuration.

That is to say, each stage "cascades" into the
next - just as a stream "cascades" over its riffles as it runs down a
slope.

Conventional theory says that up to three stages
may be connected to the same B+ supply before instability becomes a
problem.

However:

For optimum performance
each stage should be a self-contained discrete circuit - wholly independent
from those before and those after.

The usual method of decoupling is to instal a
series resistor into the B+ supply between stages. But this produces significant
voltage drop when all or any tubes draw current over the steady state
condition.

This convention also allows signal from one
stage being conducted to an adjoining stage when there is a signal
voltage difference between those stages.

So to produce effective decoupling between
stages and to prevent each stage sharing the bypass capacitors and associated
circuitry installed in adjoining stages - before or after - it is essential to
decouple effectively and completely.

To understand the OPTIMISED ELECTRON STREAM © TECHNOLOGY: SERIES DIODE DECOUPLING
concept first we need to consider the amplifier as a "system"

A "system" is defines as a series of
"processes", each having an input and an output. The ideal system is
self-adjusting or
self-correcting, by
means of a feedback loop (s).

It follows that in such a system what happens in
one part also happens in others.

In the case of an amplifier, the signal is
present on the B+ rail at each individual stage. In some cases the signal is in
phase with other stages, and in other cases the signal is out of phase with
other stages.

Therefore the B+ rail is common to all stages.
Looking at it another way (ie from the B+ line), the B+ rail is the output from
a simple
parallel mixer system - each
stage being an input to the mixer.

Clearly then the number of stages and those in
phase and those out of phase will affect the "sound" of the amplifier.

Conventional design principles regard the
junction between the plate load resistor and the immediately adjacent filter
cap as being at AC "earth" - because the filter cap is regarded as being a very
low (insignificant) impedance path for the AC in the circuit

However practical tests reveal a different
story

I submit any given electrolytic cap has
different characteristics when comparing its forward and reverse current flow.
One way to
demonstrate this is to
replace an electrolytic in the B+ rail with a non-polarised motor start cap -
the difference in tonal quality is
huge.

This proves that the junction of plate resistor
and filter (signal bypass) cap is not at AC earth at all, but at some point
above it - or at best the AC signal circuit will be affected or influenced by
the series connected bypass capacitor's inherent internal
characteristics.

Now since we are talking a VOLTAGE amplifier we
do not need to consider the CURRENT in the B+ rail but the VOLTAGE

Specifically, the TRANSIENT PEAK VOLTAGE

When a transient signal appears on the B+ rail
at any point it will appear across the entire rail - not just at the stage in
which it
originates

So if, for example, the phase-splitter B+
voltage momentarily sags, the signal from the previous stage - which does not
suffer as
much sag because it is a low
current stage - will flow to the phase-splitter stage and be mixed with it at
the filter cap

Ohm's Law applies, so wherever there is a
voltage difference you will see current flow - in whatever direction it
chooses

The purpose of the series diode in the B+ rail
is to prevent SIGNAL voltage from transferring forwards (positive
feedback)  in the
circuit whenever
a voltage difference in the B+ rail appears

It will not of course reduce negative feedback
between stages through the B+ rail

Provided a suitable diode is installed to each
and every stage in this way, this simple device prevents the AC in any
single-stage circuit from being shared by an adjoining upstream (line-side)
stage - thus ensuring all of the signal current is passed through the stage
discrete bypass (usually "electrolytic") filter capacitor.

Thus each stage can be optimised in its own
right.

9.
OPTIMISED ELECTRON STREAM © TECHNOLOGY:
SYMMETRICAL BALANCED AC
SIGNAL DRIVE SYSTEM.

This element of OPTIMISED ELECTRON STREAM © TECHNOLOGY describes the
design principle wherein the AC signal drive system circuit comprises a symmetrical balanced AC signal drive system whereby the AC
signal OUTPUT from the Driving stage circuit and the AC signal INPUT of the
Driven stage circuit must both be directly connected to a common central
axis at reference voltage potential.

That is to say, in a push-pull amplifier stage,
the mid-point central axis of the AC push-pull signal input voltage MUST be
directly connected to the mid-point central axis of the the AC push-pull signal
output voltage - ie the INPUT voltage to the driven stage.

Transformer Coupling/Driver System

In a conventional transformer coupled circuit,
the supply terminal of the single-ended driving side, or centre-tap of the
push-pull driving side is earthed via the B+ bypass cap.

In this case, the return circuit for the AC
signal is through the transformer primary back to the driver cathode via that B+
bypass cap.

Being connected to earth it follows that the
driver cathode bias resistor should be also connected to earth. In the diagram
below that would be via a centre-tap on the driver Triode AC filament
transformer.

In some Class B designs using zero bias tubes,
and in a cathode bias design, the centre-tap is usually connected to ground as
shown.

But in a fixed bias design, the centre-tap of
the transformer secondary is usually connected to the bias supply.

Thus when the centre tap of the secondary is
connected to the bias supply then the primary and secondary centre-taps do not
share a common connection - except through the bias supply earth point, which is
at positive polarity in respect to the bias voltage.

This can be seen in the following design, where
the single-ended driver is coupled to a push-pull output stage.

Note the cathode bias resistor and bypass
capacitor in the driver stage are grounded but the centre-tap of the
transformer
secondary is connected to
the negative voltage DC bias supply.

But being a transformer none of that matters,
because the primary and secondary are two completely independent and

isolated circuits.

RC Coupled Driver Stage

However when a tube RC coupled driver circuit is
used with either plate or cathode output - then the two circuits must share a
common in and out AC signal axis - or else the sound will be affected.

This is explained below.

Fig 1: Cathode Biased Output Stage

In a cathode biased amplifier the requirement
for driving and driven circuits to share a common connection is normal,
conventional design practice. A typical standard configuration is shown in Fig
1.

Notice how the central axis of the driving and
driven circuits is common. In this case the central axis is at ground
potential.

Since the circuitry in both halves of the
push-pull circuit are symmetrical and exactly equal it follows that the AC
signal voltages in both halves of the push-pull circuit are symmetrical and
exactly equal.

However this is not the case with fixed bias
amplifiers.

Fig 2 and 3: Fixed Biased Output
Stage

Fig 2 shows a conventional fixed bias
circuit.

Fig 3 shows a conventional fixed bias
circuit with the bias supply shown.

Notice how these circuits also appear to be
balanced each side of the push-pull central axis.

However these designs
are not balanced and are not symmetrical.

Further explanation is provided below.

Fig 4: OPTIMISED
ELECTRON STREAM © TECHNOLOGY Fixed Bias
Symmetrical Balanced Drive System.

Notice how this circuit appears to be the same
as Fig's 2 and 3.

However there is a subtle difference.

Notice how the Cathode circuit of the driving
stage is not connected to ground - as is conventional design practice - but to
the bias supply for the driven stage.

This is the profound difference.

Further explanation is provided below.

Fig 6: Conventional Fixed Bias
System.

Fig 6 shows the AC signal path in a conventional
fixed bias system.

Notice how the AC signal must pass through the
bias supply components - particularly the bypass electrolytic filter capacitor
(usually of the electrolytic variety). In this case, the bias supply shown is
simple, but in most designs the bias supply is complex and often comprises
chokes and other harmonic producing or refining components - in complex
inter-relationships.

Moreover, it is common practice to use half-wave
rectification in bias supplies, so the residual ripple voltage will appear as a
series of pulses in series with the grid driving circuit.

Consideration must also be given to the effects
of spurious harmonic and transient spike signals injected into the bias circuit
from the bias supply mains transformer.

Since this part of the
driving circuit is common to both halves of the push-pull drive, it follows that
the signal will ALWAYS be modified by the bias supply
characteristics.

If that was not enough to contend with, it can
also be seen that the bias supply negative DC voltage is permanently in the
signal circuit.

Furthermore, the bias voltage is offset
negatively to the centre-line axis of the signal voltage, which is nominally at
ground potential. The greater the applied bias voltage the greater the
offset.

Since the AC signal alternations in both halves
of the push-pull circuit have both positive and negative polarity, it follows
that the bias supply voltage will enhance the signal in the negative alternation
and offset the signal in the positive alternation.

This DC voltage and current source will try to
support the flow of AC signal sourced energy to the grids when grid current
flows in Classes AB1 and AB2 or B - and also Class A if gas is present in the
power tubes . However to get to the grids it must first pass through the driving
tube circuit. Since the interstage coupling capacitor will not pass DC it
follows that the DC component of grid drive power will be dissipated into the
plate resistor of the driving tube.

But along the way it can increase the cathode
bias voltage of the driving tube (which will attenuate the signal) whilst
simultaneously reducing the plate voltage (which will attenuate the signal and
reduce AC output voltage)  - thereby affecting the capability of the
driving stage to respond to the signal input.

Since the plate load resistor of the driving
tube provides an alternative return circuit (to ground) for the signal voltage
it follows that the conventional fixed bias configuration which adds DC currents
into the signal circuit includes elements guaranteeing circuit instability. When
negative loop feedback is added to such a circuit - eg from the speaker
secondary windings of the output transformer back to the driving stage - then
the result must be difficult to predict or control.

Finally, the bias supply final shunt load
resistor is in series with the output tube grid resistor and forms an integral
part of the grid to cathode circuit of the output tubes. It is common to see
relatively high values of bias supply final shunt load resistor compared with
the output tube grid resistor value - eg 47k to 100k. Sometimes complex bias
adjusting networks are also in there. It follows that the closer the (total)
bias supply resistor is to the grid resistor value then the more effect the bias
supply characteristics will have upon the AC signal behaviour.

Fig 5:  OPTIMISED ELECTRON STREAM © TECHNOLOGY

Fig 5 shows the symmetrical balanced AC signal drive system whereby the AC
signal OUTPUT from the Driving stage circuit and the AC signal INPUT of the
Driven stage circuit have a common central axis at reference voltage
potential.

In this example, the bias is set at -40
VDC.

Notice how using this system as shown in Fig 4,
the signal path circuits on both sides of the push-pull circuit are exactly
equal and balanced - in a similar manner to that of Fig 1: Cathode Bias Output
Stage.

Moreover, the signal path is direct between both
driving and driven stages and is completely independent of and therefore
unaffected by the bias supply.

Consequently neither actual bias voltage nor
bias supply characteristics will affect the signal circuit.

Note that the grid to cathode circuits of the
power tubes is the same for both Fig 5 and Fig 6 designs.

This system is very stable and provides enhanced
frequency response and reduced distortion.

It is entirely suitable for a cathode-follower
driver stage - ensure the base of the cathode follower load resistor is
connected to the bias supply as shown in Fig 5.

An added benefit for a cathode-follower driver
system is that the cathode load resistor now spans the bias voltage as well as
the B+ voltage. Consequently the cathode follower load resistor may be made
substantially larger whilst still maintaining the same plate to cathode voltage
in the tube. This can offer increased output voltage to the driven stage.

Load Resistor

Generally speaking, the greater the load
resistance/impedance to the driving stage the higher the output
capability.

However there are practical limits to this
objective since output tube grid circuit resistance must be held within
manufacturers' published specifications.

To maximise the driver stage load whilst
minimising driven stage grid circuit resistance the RATIO of driver
stage load plate resistor to driven stage grid circuit load resistor should be
held within the range of about 1:1 to 1:2.

Most of us have a junk box
stock of perfectly good tetrodes, pentodes or beam power tubes just waiting to
be used - so why not experiment and prove the benefits of OPTIMISED
ELECTRON STREAM © TECHNOLOGY to
yourself.

There is no restriction or
cost imposition upon the home hobbyist constructor to using these concepts - the
only restriction is on commercial exploitation - so if you do not like it do not
do it.

If you want your hi-fi to
improve its performance at minimal cost to you then experiment. The concepts
presented here do work and cost very little to implement.

However to those who say
that a product is only as good as what you pay for it, then these
concepts are of no value to you because they are free. You would be wiser to
spend a hundred grand on a commercial system and feel better. While you are so
doing, ask the manufacturer to justify the circuit design and component choices
to you - ie why the design is what it is and not some other alternative
approach. That is "why is it better?"

To those who consider
relating these concepts to RF applications - this is an audio focused site -
experiment at your own risk. It may be safer to stay with the tried and true -
waste a little power and live with it.

However RCA, in their Transmitting Tube Manual
TT4, say at Page 44 -" The only restrictions on tube operating values are those
imposed by the published maximum ratings." The OPTIMISED ELECTRON STREAM © TECHNOLOGY design
concepts presented above should enable designers to withdraw operating
conditions back into specification whilst improving performance.

Finally, an OES tube
amplifier can only and will always sound like a tube amplifier. The OES concept
is limited to and by the vacuum tubes it relates to.

HAPPY
CONSTRUCTING!!

MAY YOUR PROJECT BE A
SUCCESS!!

REMEMBER:

- ALWAYS TAKE CARE WHEN
WORKING WITH HIGH-VOLTAGE -

DEATH IS
PERMANENT!!

©
NOTICE:

INTELLECTUAL PROPERTY COPYRIGHT © D.R.GRIMWOOD 2002 - ALL RIGHTS
RESERVED.

Intellectual property in the applied engineering concepts expressed in
this paper remains exclusively with the author.

IMPORTANT
NOTICE

THE AUTHOR MAKES
NO CLAIM WHATSOEVER AS TO THE VALIDITY OR ACCURACY OF ANY STATEMENT,
INFORMATION OR OPINION CONTAINED IN THESE PAGES AND NO LIABILITY WILL BE
ACCEPTED FOR ANY ERROR OR OMISSION OF ANY KIND
WHATSOEVER.

PLEASE NOTE NO
WARRANTY IS EXPRESSED OR IMPLIED AS TO THE WORKABILITY OR PERFORMANCE OF
DESIGN INFORMATION DESCRIBED
HEREIN.

For suggestions, critique
or discussion re this page contact:

DENNIS
GRIMWOOD

Email:   contact

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This page last amended 25 April 2016