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Class Distinction in Audio Amplifiers

A discussion of design problems and how to overcome
them

by J. L. Linsley Hood
(1)

(Wireless World, June 1970)

Since the publication of "Simple Class A Amplifier" the author has received
numerous letters asking whether it would be feasible to increase the power
output to 15W, or even 20W, to provide a greater reserve for use with
inefficient loudspeaker systems.

Whilst it would be possible, the problems associated with increased heat
dissipation and the provision of suitable power supplies makes this
unattractive. In view of the low average power required for normal listening,
the question inevitably arose whether it would be practicable to design an
output stage which would operate in class A with an inherently low level of
high order distortion up to a watt or two, but progress further into class B
operation if and when higher powers were momentarily demanded.

There are, unfortunately, a number of snags with the class B operation of
transistor output stages, to which the answers are not fully known.

It was pointed out some years ago, by Bailey (2) and others, that the use of quasi
complementary symmetry in such output stages led to an increase in high-order
harmonic distortion, associated with the non-linearities in the crossover
characteristics at low volume levels, and although the level of total harmonic
distortion at maximum power output could be quite low, the distortion content at
typical listening levels could be many times greater than this, and would also
be of an audibly objectionable type.

A number of schemes have been proposed to overcome this problem, including
the use of full complementary symmetry (2 3 4), and
various methods of ensuring that there are an equivalent number of forward
biased junctions in each limb have been described (5 6),
including the ingenious semi-complementary triples arrangement used in the
"Quad" amplifier (7).

However, in the author's experience, some class B transistor amplifiers -
including those employing full symmetry, which is presumed to eliminate the
major fundamental snags of this type of operation - having an impeccable
performance on paper, did not have the tonal quality which had been expected.
Since harmonic distortion at both high and low power levels had been found to
be well below the level at which audible effects might reasonably be expected
in some of the designs tested, it seemed more probable that the audible
ill-effects were due either to transient instabilities associated with
loudspeaker loads -perhaps related to changes in the reactance of the
base-emitter junction at the current cut-off point - or to high-frequency
crossover type distortion arising from hole-storage effects. Hole-storage
depends on the presence of holes produced when current flows in a semiconductor
- even though the current is due to majority carriers (electron flow). The
greater the current the greater the number of holes and the worse the problems
of hole storage.

Hole-storage phenomena

The expected result of hole storage in the base region of a transistor,
following the attempted termination of a high emitter collector current, is
that the transistor remains in a conducting state after the forward base bias
has been removed. This has the effect, amongst other things, that the normal
crossover discontinuity shown in Fig. 1(a) becomes displaced from the mid-point
of the transfer waveform as the frequency is increased, as shown in Fig.
1(b).

These waveforms were generated in a simple complementary pair
emitter-follower circuit, without additional negative feedback, driving a
resistive load. (In order to assist its display the crossover effect was
deliberately exaggerated by the use of an inadequate quiescent current.)
Provided that the peak currents flowing through the transistors are small, this
effect is innocuous. However, if the peak currents are increased, by reducing
the load resistance, the crossover waveform rapidly deteriorates as shown in
Fig. 1(c), and increasing the forward bias to give a more suitable quiescent
current has little effect in removing this prominent notch, until the forward
bias is almost equivalent to that of class A operation.

It is known from experience that these effects can be minimized by the use of
transistors with good high-frequency characteristics and low-impedance
base-emitter return paths. A low-impedance driver stage will also be effective
provided that it does not become cut off (as in the case of the Darlington
pair) when the input signal reverses polarity.

The effect of reducing the driver circuit impedance from 2000ohm to 100ohm is
shown in Fig. 1(d).

The lack of effective symmetry between the upper n-p-n device and the lower
p-n-p is also shown in Fig. 1(c). This effective asymmetry is reduced if the
source impedance is reduced.

Fig. 1. Crossover distortion in
a class B stage employing transistors with an fT
of about 2MHz. (a) Low frequency sine wave at 10mA. (b) High frequency sine wave
showing the effect of hole storage on the crossover discontinuity under light
load conditions. (c) Influence of hole storage and n-p-n/p-n-p asymmetry under
high current conditions at 200kHz. (d) Improvement of conditions in (c) by
reducing source impedance.

It was noted that this effect did not become apparent, even under high
emitter current conditions, until the operating frequency approached 0.05
fT. At 0.1 fT, the problem was severe and this argues that
the occurrence of high transient currents - which  may  arise
with  certain  loudspeaker systems - and high driver stage output
impedances, is most undesirable unless the highest frequency components of the
waveform are low in relation to the transition frequency of the output
transistors. With the availability of power transistors having transition
frequencies of the order of 4MHz (such as the MJ480/490 series) it is unlikely
that hole-storage phenomena will be troublesome at the rates-of-change of signal
voltage likely to be encountered in audio amplifier practice so long as the
driver stage does not leave the output transistor base open-circuited on
cut-off. However, the use of a driver output, or base circuit, impedance not in
excess of a few hundred ohms appears prudent. With earlier designs using
germanium diffused junction power output transistors, which usually have very
poor h.f. performance, this problem could be important, and Dinsdale has
referred to a "subjective audible improvement" resulting from the replacement of
low transition frequency output transistors with types having better h.f.
characteristics.

Transient instabilities on loudspeaker loads

Phase-angle measurements made with a variable frequency sine wave input, from
a high impedance source, reveal that even a simple single-unit loudspeaker can
present quite complex characteristics. The reactance - which is normally
inductive - changes rapidly, and sometimes even becomes capacitive, at
frequencies in proximity to cone and structure resonances.

In general, the characteristics of most of the common designs of transistor
power amplifiers are such that instability problems do not arise with inductive
loads, and the inclusion of a small choke, of a few microhenries inductance, in
the speaker output lead is a well known technique for avoiding instabilities
under adverse load conditions. However, capacitive loads can frequently impair
the stability margins of the feedback loop, and it is in this respect that

the reactive characteristics of the loudspeaker load are most significant
Since it was suspected that the region of the output waveform where this might
arise most readily was that at which the output transistors were being driven
from the conducting to the cut-off state, an input waveform which provided a
transient of controllable steepness (by varying the input amplitude), but
arrested at the mid-point, was provided by the circuit of Fig. 2.

Fig. 2. Circuit for generating
the test waveform shown in Fig.3.

The waveform generated by this device is shown in Fig. 3 and the result of
introducing such a waveform into an amplifier of poor stability margins,
coupled to a resistive load shunted by an appropriate value of capacitance is
shown in Fig. 4(a). (The broadening of the oscilloscope trace in the horizontal
regions at the mid-point of the waveform was due to inadequately recorded h.s.
oscillation.)

Fig. 3. Test waveform for
providing arrested transient input.

The output waveform obtainable from a design with better stability margins
and improved bandwidth is shown in Fig. 4(b). In both cases the magnitude of
the input signal was adjusted so that clipping occurred on both negative- and
positive-going peaks.

Since the h.f. instability shown in Fig. 4(a) - which did not occur in the
absence of a large input signal, and which required a particular range of shunt
capacitance to provoke it at all - also occurred on parts of the waveform
preceding the arrested transient, it was concluded that the change in reactance
of the base-emitter junction at cut-off or switch-on, was not a major cause of
the transient induced instability observed in this particular design.

Fig. 4. Amplifier performance
using 10kHz test waveform. (a) Response of amplifier showing inadequate
stability with reactive load. (b) Response of improved amplifier with reactive
load.

Square-wave performance and tonal quality

In view of the fact that a loudspeaker system can present a reactive load, of
a type which is found in certain circumstances to cause signal induced
instability, and since this instability could be provoked by a square-wave
input into an amplifier with a suitable reactive load, a series of tests and
comparative listening trials was conducted to determine whether there was any
audible relationship between the two. In the event, it was found, beyond doubt,
that an amplifier system which did not show any sign of instability over the
range of load shunt capacitances up to, say, 0.33uF had a better tonal quality
on even a simple loudspeaker system than one in which some shunt capacitor
value could cause h.f. oscillation. Moreover, in a more complex loudspeaker
system, with a crossover network and high-frequency capacitively coupled
"tweeter", it was possible to hear the difference between systems which would,
in the lab., with some RCload combination, give a square-wave response
such as that of Fig. 5(a) and those which had a response like that shown in Fig.
5(b). No positive distinction could be drawn in listening trials between a
system giving a waveform such as Fig. 5(b) and one in which a square-wave input
could produce a single overshoot "spike".

Since the frequency of the "ring" waveform in Fig. 5(a) is well beyond the
upper limits of the audible spectrum, it is clear that it is not this of itself
which produces the undesired sound quality, but rather that this type of
behaviour is symptomatic of a different and more objectionable effect when the
amplifier is used with a loudspeaker load.

Fig. 5. Amplifier response
driving a reactive load (15ohm, 0.47uF) with a 10kHz square wave.

(a) The ringing gives evidence
of instability. (b) No transient ring indicates better stability.

The conclusions which have been drawn from this series of experiments are
these: (1) that it is desirable to employ output power transistors in which the
transition frequency is at least ten times higher than the highest signal
frequency component which is passed to the amplifier from preceding stages; (2)
that it is preferable to drive the output transistors from a source which has a
low impedance over the whole signal voltage swing, or at least to provide a
reasonably low-resistance base-emitter current path; and (3) that the
phase/frequency characteristics of the feedback loop should be such that a
square-wave output devoid of overshoots is obtained when the amplifier is bench
tested with a wide range of shunt capacitance values in an RCdummy load.
This latter requirement probably implies either a fairly limited number of
stages within the feedback loop or a relatively restricted h.f. bandwidth.

When these requirements had been met, and when the harmonic distortion levels
over the range 40mW up to the maximum rated power output were of a suitably low
level, there was no audible difference, in the most careful listening trials,
between several different designs. However, it is difficult in class B systems
to obtain the desired low level of harmonic distortion at low signal levels
without the use of substantial amounts of negative feedback, and this leads to a
worsening of the amplifier response to signals containing transients.

The use of a class AB system, if the problems in maintaining the correct
forward bias level can be solved satisfactorily, should facilitate the
attainment of these desired standards, particularly if the h.f.
negative-feedback loop can be made fairly simple.

Next month full details will be given of a 15-20W class AB amplifier with the
following characteristics:-

Power output: 15W into 15ohm, or 18W into 8ohm (20W with modified
output circuit component values.)

Bandwidth: 10Hz-100kHz +/- 0.5dB at 2V output; 20Hz-50kHz +/- 1.0dB at
maximum power output.

Output impedance: 0.03ohm (at 1kHz).

Total harmonic distortion: 0.02% at 15W/15ohm or 18W/8ohm; less than
0.02% at all power levels below maximum output.

Intermodulatlon distortion: Less than 0.1% at 10W (12.3V r.m.s. into
15ohm) and 70Hz, and at 1V r.m.s. at 10kHz.

Square-wave transfer distortion: Less than 0.2% at 10kHz.

REFERENCES

1.  Linsley Hood, J. L., "Simple Class A Amplifier", Wireless World,
April 1969.

2.  Bailey, A. R., "30-watt High Fidelity Amplifier", Wireless World,
May 1968.

3.  Williamson, R., Hi-Fi News,Feb.1969, pp. 320-329.

4.  Hardcastle, I., and Lane, B., "Low-cost 15-W Amplifier", Wireless
World, Oct. 1969.

5.  Shaw, I. M., "Quasi-complementary Output Stage Modification",
Wireless World, June 1969.

6.  Baxandall, P. J., "Letters to the Editor", Wireless World,
Sept.1969.

7.  "Low Distortion Class B Output", Wireless World,April
1968.

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