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Maxim > Design
support > App notes
> Audio
Circuits > APP 3878

Aug 18, 2006

Keywords: class D, EMC, audio, amplifier,
filterless, modulation, pwm
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APPLICATION NOTE 3878

Reduce EMI from Class D Amplifiers Using New
Modulation Techniques and Filter Architectures

Abstract: Class
D amplifiers have always offered better efficiency than Class
AB amplifiers for low-power applications. This inherent
advantage, however, rarely outweighed the traditional
disadvantages of Class D, which were higher cost, reduced
audio performance, and the need for output filtering. Recent
advances in Class D technology, however, have brought Class D
costs and audio performance up to equal Class AB. Some novel
Class D output modulation schemes, moreover, now ease the EMI
burden in many applications.

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Introduction
The state of the art in Class D
amplification has progressed rapidly in the past few years,
most noticeably for lower power applications that require less
than 50W per channel. For these low-power applications, Class
D has an inherent efficiency advantage over the traditional Class
AB amplifier, because the Class D output stages are always
on or off, with no intermediate bias stage necessary.
Historically, however, this efficiency advantage never held
widespread appeal for designers, because Class D also had
notable disadvantages: higher parts cost, poor audio
performance (compared to Class AB), and the need for output
filtering.

Two major factors have reversed this design
trend in recent years, making Class D more appealing in a wade
range of applications.

First, there is market
need. Two rapidly growing end-equipment markets benefit
from Class D amplification in different ways; cell phones and
LCD
flat-panel displays. With cell phones, both the speakerphone
and PTT (Push-to-Talk) modes benefit from Class D's higher
efficiency, which provides longer battery life. The growth of
LCD flat-panel displays generated a need for 'cool running'
electronics, as the display color contrast suffers under
elevated operating temperatures. As Class D efficiency means
that less power is dissipated in the drive electronics, the
LCD flat-panel displays operate cooler and with better looking
pictures.

The second factor to affect use of Class D is
the technology itself. Driven by market needs, several
manufacturers now offer improved Class D technology with costs
more equitable and audio performance on a par with Class AB.
Some novel Class D output modulation schemes, moreover, now
ease the EMI
burden in many applications.

Some of the newer Class D
designs, although derived from older PWM-style architectures,
incorporate sophisticated modulation techniques that achieve
filterless operation for lower power systems. Efficiency
claims can be verified on the bench, but some designers
suspect that products based on these new techniques will be
rife with EMC/RFI compatibility problems. In reality, the use
of effective PCB
layouts and short runs of speaker cable
can ensure sufficiently low radiated EMI to pass the
applicable FCC or CE standards.

Application Challenges
The physical layout in some
applications necessitates long speaker cables, for which RF
emissions must be more tightly controlled because the speaker
cables act as antennae. In effect, the longer the speaker
cable, the lower the frequency at which it acts efficiently as
an antenna. Meanwhile, other applications require EMI
emissions below that of CE/FCC, perhaps to meet automotive
specifications or where interference with other circuitry at
lower frequencies must be avoided. With such disparate
requirements, these applications were historically difficult
to overcome.

A flat-panel TV is an obvious example of
the above application dilemma. With speakers typically arrayed
at the outer edges of the device, long speaker cables are hard
to avoid. If analog
video signals are present, simply meeting FCC or CE RF
emissions may not be sufficient; (these limits are specified
from 30MHz upwards) ad suppression of the switching
fundamental may be necessary to avoid interference effects
with the video signal. If you need to use the traditional LC
filters that operate effectively with older PWM amplifiers,
you should analyze them to ensure that they are effective in
suppressing the high-frequency switching transients produced
by the latest amplifiers.

PWM-Based Class D Amplifiers
Traditional Class D
amplifiers are usually based on the principal of pulse-width
modulation (PWM). Their outputs can be configured either
single-ended or as a fully differential bridge-tied
load (BTL).
Figure 1 shows the output waveforms typical of a BTL,
PWM-based Class D amplifier. The fast switching times and
nearly rail-to-rail swings give this type of amplifier very
high efficiency. However, the wide-output spectrum implied by
those same parameters can lead to high-frequency RF emissions
and interference. Consequently this design approach usually
includes output filters to suppress the unwanted RF
effects.

Figure
1. Waveforms of the traditional pulse-width modulation (PWM)
scheme.

As Figure 1 shows, the mirror-image
waveforms assert very little common-mode (CM) signal on the
speaker or cables (lower trace), provided that the waveforms
of the inverting and noninverting output devices are well
matched. Note that a 50% duty cycle represents a zero input
signal (idle). You can, therefore, design a differential
lowpass filter that attenuates high-frequency content in the
waveforms (due to the rapid switching), but preserves the low
frequencies intended for the loudspeaker.

New Modulation Techniques
With interest in Class D
amplifiers growing, several manufacturers recently introduced
new modulation schemes that independently control the two
halves of the H-bridge.
These modulation schemes offer two key advantages:

Virtually no differential switching is present across
the load for very low audio signals and at idle. This
improves quiescent
current
consumption over traditional PWM designs.

The minimal pulse, CM switching helps make the turn-on
and turn-off transients lower in level. The DC idle level
(post filtering) on each of the BTL 'legs' is close to GND.
Any mismatch, therefore, from filter components or stray
capacitance (which can cause audible click-and-pop
when the amp is enabled or disabled) is minimized.

Clearly, there are some advantages to the new
techniques, but the amplifier outputs are no longer mirror
images of each other. The waveforms shown in Figure 2
(representing the MAX9704 stereo Class D amplifier) have a
high level of common-mode content.

Figure
2. Modulation scheme for Maxim's MAX9704 stereo Class D
amplifier.

The output-filter requirements for these
Class D amplifiers differ, from those requirements for an
amplifier with the traditional differential and complementary
PWM outputs. Compared with PWM, the MAX9704's modulation
scheme includes a high level of CM signal, and any
output-filter design should take that into account. A
traditional differential filter topology can give poor
results, as the following example shows.

Figure
3a depicts a traditional LC, PWM, Class D output filter,
implemented with ideal values. For simplification, the speaker
load is represented as a pure 8Ω resistance
and the inductors' DC resistance is assumed to be negligible.
The problem can be highlighted with some straightforward SPICE
simulations. Figure 3b shows the response of the Figure
3a filter driven by a differential input signal. Each output
node (FILT1, FILT2) is plotted with respect to GND. The values
chosen give a second-order slope above 30kHz and a
well-controlled transition. Group delay is flat across the
audio band
at approximately 4µs.

More
detailed image (PDF, 654.5kB)
Figure 3. The response
of a traditional differential-mode passive LC filter
(a) is different for differential-input signals
(b) and common-mode signals
(c).

Figure 3c shows the same filter
output driven with a CM signal. Again, each output is plotted
with respect to GND. The result (note the shifted Y axis!) is
heavily peaked and obviously very underdamped. That result is
easily understood by considering how the filter appears to a
CM signal (Figure 4). Because the simulation provides
ideally matched inductors and capacitors, the differential
signal across the resistive load is zero, and subsequently has
no damping effect on the LC components. L1 interacts with C1
(as does L2 with C3) to give the peaked response. In the time
domain (not shown) this condition would indicate heavy
overshoot and ringing. Note also that C2 makes a zero
contribution when driven by CM. This means that the filter's
cut-off frequency (or more accurately stated in this case, its
resonant frequency) is higher than that of the differential
case.

Figure
4. The traditional LC filter of Figure 3a looks like this to a
common-mode signal.

At this point you can ask, is
this a problem? Probably not, if the output spectrum has zero
common-mode energy at that frequency. If, however, the peaking
frequency coincides with the Class D switching frequency,
large voltage-output excursions can appear at the speaker and
the cabling. Further, the MAX9704 in its
spread-spectrum-modulation (SSM) operating mode can excite the
underdamped filter by producing appreciable noise energy above
the audio band. Spread
Spectrum Mode is a pin-selectable option in which the
high-frequency switching energy is 'whitened' and lowered in
amplitude by randomizing the switching period on a
cycle-to-cycle basis. This spread-spectrum approach also eases
EMI compliance in a filterless design.

Possible Solutions to the Underdamped CM Response
One
solution to this CM problem is to preserve the basic
architecture of Figure 3a, but add damping elements that
suppress the highly resonant common modes. Figure 5a
shows the addition of two series RC elements from each output
node to GND. If efficiency is not important in your
application, you can simply add resistors to GND, but
capacitors C4 and C5 help to minimize excessive power
dissipation in R1 and R2.

The values of C4 and C5
impose a tradeoff; they must be large enough to allow R1 and
R2 to damp out the peaking, but small enough to minimize the
power loss at high audio frequencies (usually up to 20kHz).
This tradeoff is made easier if the CM cutoff frequency is
much higher than the differential-mode frequency, a condition
implemented by increasing the ratio of C2 to C1 and C3. By
increasing the CM cutoff frequency, C4 and C5 can be smaller
and R1 and R2 larger, thereby minimizing the audio-frequency
power loss into R1 and R2. Pushing the CM cutoff frequency too
high allows more CM on the cables> You must, therefore,
determine a reasonable limit in the ratio between the
differential and CM -3dB points. For this filter, we adopted
1:5 for that ratio.

More
detailed image (PDF, 693.4kB)
Figure 5. Adding an RC
network on each output of the traditional LC filter (a)
improves the response for differential signals (b) and
CM signals (c).

Figure 5b shows the
filter of Figure 5a driven differentially, and Figure
5c shows the response when driven by CM. Note now the
higher frequency CM cutoff in Figure 5c (-3dB at around 110KHz
vs. 28kHz for the differential case), with gentle but
well-controlled peaking. This cutoff is well above the highest
audio frequencies (and below the Class D switching-frequency
fundamental), so it should be of little
consequence.

Some applications with low switching
frequencies (200kHz to 300kHz) will not work well with the
approach shown in Figure 5c. For those products you may need
to adopt other methods and topologies. The MAX9704 stereo,
Class D amplifier (Figure 6) gives best results when
set for Fixed Frequency Mode (FFM) operation at 940kHz (FS1 =
low, FS2 = high). FFM in the MAX9704 sets the switching period
at a constant value, pin selectable among three values, to
suit a given application.

Figure
6. Typical application circuit for the MAX9704, stereo Class D
power amplifier.

Figure 7 and Figure
8 illustrate the time-domain performance for the Figure 5
filter when driven by the MAX9704. A resistive 8Ω load was
used in both cases. Figure 7 shows the FILT1 and FILT2 nodes
overlaid (top traces), and the resulting 1kHz differential
output waveform (lower trace). Noise on the upper traces is
the residual of the output switching after filtering. (Supply
voltage is 15V.) Figure 8 shows a detail on the trace of
Figure 7. Note that the ripple, mostly from the 940kHz
switching frequency, appears as CM on both channels. Note also
the absence of higher harmonics, which shows the effective
suppression of EMI frequencies. (Radiated-EMI measurements
usually start above 30MHz.)

Figure
7. Driving the Figure 5a circuit with a MAX9704 produces
waveforms at FILT1 and FILT2 (overlayed in the top trace) that
combine as a differential output (bottom
trace).

Figure
8. The top traces show residual ripple voltage at the outputs
of Figure 5a, where the main ripple component is the
switching-frequency fundamental (940kHz, in this case). The
filter's second-order slope above that frequency heavily
suppresses any higher harmonics. The ripple is almost all CM
(lower trace).

Further Notes
The filter designs in this article all
assume a resistive load of 8Ω. Voice-coil inductance causes
the impedance
of most wide-range moving-coil loudspeakers to rise above
~20kHz. That property makes efficient filterless operation
possible, but you must account for the rising impedance when
optimizing the component values for any additional EMI output
filtering.

When attempting to evaluate and characterize
the performance of a Class D amplifier, audio designers also
need to use filtering in the lab for part selection &
evaluation purposes. Even if the end product can pass EMC
tests without filters, the amplifier characterization can pose
problems. Many audio analyzers intended for measuring THD+N or
amplitude response from conventional audio amplifiers can give
false results when driven by a filterless Class D amplifier.
While the Figure 5 circuit is also useful on the test bench
(when loaded correctly with an 8Ω resistor), note that the
33µH inductors can introduce nonlinearities that can limit THD
measurements. Air-cored components give best results, but
their size can limit usage in actual products!

A
similar article appeared in the February 16, 2006 edition of
Electronic Design.

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Download,
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APP 3878: Aug 18, 2006

AN3878, AN 3878, APP3878,
Appnote3878, Appnote 3878

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