Design Feature:
May 25, 1995

Class D amplifiers provide high
efficiency for audio systems

Jeffrey D Sherman,

Harris Semiconductor

Advances in
MOSFET technology and integrated half- and full-bridge predrivers now make
class D amplifiers a practical alternative to linear amplifiers in many
applications. The biggest benefit compared to more traditional amplifier
topologies is high efficiency.

Many engineers are familiar with motor-control circuits that
pulse-width-modulate a voltage to control both the direction and speed of a dc
motor. You can extend this concept to an audio amplifier by replacing the motor
load with a lowpass filter and a speaker and by using an audio input as the
control signal. Such amplifiers are class D switching amplifiers, which are
analogous to conventional class A, B, AB, and C circuits, in which each
switching element is in its linear mode of conduction for a large percentage of
each cycle of an audio input signal. This linear operation reduces efficiency
to about 60% in the typical class AB amplifier and necessitates the use of
large heat sinks.

Alternatively, the switching elements of a class D amplifier are either cut
off or in saturation most of the time, allowing high efficiencies. The high
efficiency translates into reduced heat sinking, smaller size, and lighter
weight. Also, class D amplifiers do not suffer from crossover distortion.

An evaluation-board design shows efficiencies of 94% and THD-plus-noise
(THD+N) performance that ranges from less than 1 to 2.8%. Although the THD+N
curves for this board's design are not as good as what you can obtain with a
high-performance class A or class AB amplifier, the curves are more than
adequate in many applications. For instance, automotive manufacturers typically
require amplifiers in car stereos to have distortion below 2% at lower power
levels and below 5% at peak power. Also, optimizing some of the supporting
circuits should result in even higher performance than that of this evaluation
board.

The concept of a class D switching amplifier has been around for about 50
years. Early attempts to develop switching amplifiers with vacuum tubes were
limited by the tubes' large voltage drops and low current capabilities, which
reduced the amplifiers' efficiencies and limited their output power. In the
late 1960s, bipolar transistors became a practical alternative to vacuum tubes
and allowed the implementation of switching amplifiers with very high
efficiencies at low frequencies.

However, an audio switching amplifier requires high-frequency operation,
which is generally equal to at least four or five times the bandwidth of the
20-kHz audio spectrum. Higher frequency operation makes it easier to design the
filter that removes the carrier frequency before the audio signal drives the
speaker. Using bipolar transistors at the required frequency of 80 kHz or
greater results in excessive switching losses that eliminate the class D
amplifier's efficiency advantages.

In
the 1980s, MOSFETs became available that could meet both the switching-speed
and conduction-loss requirements to effectively implement class D amplifiers.
The first switching amplifiers using MOSFETs incorporated electrically isolated
drivers to allow the use of N-channel devices. N-channel MOSFETs yield more
efficient designs; these MOSFETs have approximately one-third the conduction
losses of their P-channel counterparts. However, the isolated drive circuits
were complex and limited the use of switching amplifiers.

Fabrication processes that allow the integration of high-voltage and logic
circuits on one die made possible the first commercially available integrated
predrivers for class D amplifiers. For example, the Harris HIP4080 provides
monolithic isolation and allows the use of high-side N-channel MOSFETs through
the use of a bootstrap circuit and an integrated charge pump. The chip
integrates a PWM comparator for modulation. This full-bridge driver can switch
MOSFETs at frequencies above 1 MHz, reducing the size of magnetics and
simplifying the filter design.

A
conventional push-pull (class AB) linear amplifier (Fig
1) modulates load power by continuously varying conduction through its pass
elements during most, if not all, of the conduction cycle. Q1
conducts during the positive half-cycle, and Q2 conducts during the
negative. During each half-cycle, the conducting transistor operates in its
linear region. The transistor must supply the required current to the load
while reducing the voltage between the supply and the load. The power
dissipated in the transistor, which equals (VBUS-VLOAD)×ILOAD,
is wasted in the form of heat. Generally, you must use large heat sinks to
prevent the output stage from overheating.

The class AB amplifier uses a bleeder circuit to reduce crossover
distortion, which occurs during the zero-crossing of the input signal when
neither transistor is on (when the input signal is below the VBE of
either transistor). The bleeder circuit biases both transistors on during
crossover, but the circuit draws current that further reduces the efficiency of
the amplifier.

Ballast resistors in class AB designs prevent the transistors from going
into thermal runaway. Bipolar transistors are at risk because their VBEs
have negative temperature coefficients. Usually, you must mount the diodes and
transistors on the same heat sink to ensure that the VBEs track. The
common heat sink helps minimize crossover distortion over temperature.

Class D audio amplifiers

Class D amplifiers convert the audio signal into high-frequency pulses that
vary in width with the audio signal's amplitude. The varying-width pulses
switch the power-output transistors at a fixed frequency. A lowpass filter
converts the pulses back into an amplified audio signal that drives the
speakers. This design approach produces an amplifier with better than 90%
efficiency and that is more complex than its linear counterpart.

The
amplifier requires an integrator, a duty-cycle modulator, a switch predrive
circuit, and an output filter. The half-bridge class D amplifier using
constant-frequency, duty-cycle modulation (Fig
2), sums the square-wave output of the switching power transistors with the
audio input to provide negative feedback. You cannot take the feedback after
the lowpass filter unless you use a complicated compensation network to handle
the phase shift that the filter introduces. A two-pole filter, for example,
would introduce a 180° phase shift, which would cause the circuit to oscillate.

The
square-wave output is synchronous with the audio input, but you must remove the
carrier. The integrator sums the two signals and simulates the effect of the
output filter. The circuit feeds the resultant error signal into the duty-cycle
modulator, which comprises a comparator and a triangle-wave generator (Fig
3). Then, the circuit compares the triangle wave to the error signal to
produce the modulated output.

The modulated output is a square wave whose duty cycle is proportional to
the input signal. In this half-bridge circuit, this output drives the upper and
lower power switches in antiphase; the circuit always drives one switch into
saturation while it cuts the other off. The square wave causes the switches to
change state as fast as possible, given the technology used to implement the
switch. Fast switching limits the time that the switches spend in the linear
operating region, thereby increasing efficiency and reducing heat generation.
The combination of switching and conduction losses defines the upper bound of
the amplifier's efficiency.

The circuit filters out the high-frequency square wave that the power
switches generate, leaving only the amplified audio signal. This signal then
drives a ground-referenced speaker load.

Fig 4 shows a simplified schematic of
a class D audio amplifier using a full-bridge configuration. If you use a
full-bridge design, in which the circuit develops a peak-to-peak load voltage
equal to twice the supply voltage, you need only a unipolar supply instead of
the bipolar supply required for the half-bridge design. The full-bridge circuit
operation is similar to that of the half-bridge design, except the full-bridge
design uses four MOSFETs instead of two, uses a differential lowpass filter in
the feedback network to support the floating load, and requires two lowpass
filters to attenuate the carrier.

Four N-channel MOSFETs switch high currents into a low-impedance load. These
MOSFETs turn on in pairs, one high-side device with the opposite low-side
device. To avoid shoot-through, the circuit ensures a dead time between turning
off one pair of MOSFETs and turning on the other pair.

MOSFETs in a full bridge need to withstand only half the voltage they would
see in a half-bridge circuit. This lower voltage requirement allows you to
minimize conduction losses, even though two devices are in the conduction path.
Given a fixed die area, two MOSFETs in series have a lower total RDS(ON)
than does one MOSFET with twice the drain-source breakdown voltage rating (BVDSS)
because RDS(ON) increases nonlinearly with higher drain-to-source
breakdown voltage.

Base your MOSFET selection on the following criteria: peak voltage and
current requirements, body-diode reverse-recovery time, switching losses, and
conduction losses. The peak voltage and current determine the required ratings
that the MOSFET must be able to sustain. The body-diode recovery time,
switching losses, conduction losses, and output-filter losses determine the
output stage's efficiency. You can calculate peak voltage and current using the
following equations:

For example, if you require an output power of 100W into an 8(ohm) speaker,
VPEAK and IPEAK would equal 40V and 5A, respectively.
Therefore, the MOSFETs must have a BVDSS above 50V (using a guard
band of 25%) and a rated current above 5A.

Select a MOSFET with a body diode that has a low reverse-recovery time to
reduce the losses that occur when the MOSFETs switch off and on. The current
state-of-the-art reverse-recovery time is 100 nsec. You can minimize switching
losses by choosing a low switching frequency, a MOSFET with low gate-to-source
capacitance, and a driver with high drive capability. In selecting an operating
frequency, weigh the advantages of high carrier frequencies against increased
switching losses and the increase in radiated EMI/RFI. The lower the switching
frequency, the harder it is to filter the carrier from the audio signal.

If you effectively manage these losses, the conduction losses become the
dominant factor in the amplifier's efficiency. In other words, choosing a MOSFET
with a lower RDS(ON) increases the amplifier's efficiency. For
example, a MOSFET with an RDS(ON) of 200 m(ohm) would reduce the
efficiency 5% from ideal. This number results from

multiplying by 2 in the numerator because of the full-bridge topology.
However, a lower RDS(ON) of 80 m(ohm) would reduce the efficiency
only 2%. That is, you are simply trading silicon for efficiency.

Fig 5 shows the internal circuitry of
Fig
4's feedback-network block. In this circuit, the differential lowpass
filter comprising IC1C and associated components attenuates the
output of the half-bridge by a factor of 11. Buffer amplifier IC1B
ac-couples the audio input. An integrating error amplifier, IC1A,
sums the filtered feedback signal and the buffered audio input. The circuit
feeds the error signal into the inverting input of the comparator in Fig
4's driver IC, which compares the error signal with a sawtooth waveform to
generate the PWM signal. A simple triangle-wave oscillator generates the
sawtooth waveform.

Another feedback signal in Fig
4 comes from the current-sense resistor on the ground leg of the
half-bridge. The sense resistor should have as low a value as possible to
reduce its contribution to conduction losses. Simple circuitry in the
overcurrent-shutdown and reset-logic block compares the current-sense output to
a voltage reference. If the current rises above the maximum rated current of
the MOSFETs, the voltage across the current-sense resistor exceeds the
reference, causing a comparator to change state and disable the output drivers.
You could also use the current-sense resistor to implement current limiting.
This would protect the MOSFETs, but the amplifier's output would distort during
an overcurrent condition.

The
class D amplifier uses two Butterworth filters (Fig
6) to supply the amplified audio signal to the floating load. Butterworth
filters provide flat response in the passband, an important criterion in an
audio system because a flat response improves dynamic performance. Each filter
in Fig
6 has four poles with 30-kHz cutoff frequencies. The clock rejection of
each is 74 dB at 250 kHz, which is the carrier frequency for this
evaluation-board circuit. You can improve the rejection by adding poles or
lowering the cutoff frequency. An RC filter across the load provides impedance
stabilization at high frequencies. Stabilization is necessary because the
speaker becomes inductive as frequency increases, and a Butterworth filter
requires a resistive load.

When
driving 100W into a 4(ohm) load, this amplifier has a THD+N across the audio
spectrum of less than 1% to 8 kHz (Fig
7a). Above 8 kHz, THD+N increases to about 2.8% before rolling off. THD
increases because the amplifier introduces more nonlinearity as the audio input
frequency rises. As the frequency approaches the high end of the audio spectrum
(above 12 kHz), THD rolls off because the output filters begin to attenuate the
harmonics.

However, distortion at lower power levels-that is, performance at less than
peak output-is a better measure of an audio amplifier. An audio amplifier
delivers average power that is typically a fraction of its maximum output
power, depending on the dynamic nature of the music. At a 10W output level, this
class D amplifier's THD+N is less than 1.2% over all the frequencies of
interest (Fig
7b). THD increases at higher power levels because the amplifier introduces
nonlinearities as the output waveform approaches clipping.

THD is primarily a function of filter selection and the feedback network,
and you can further reduce distortion to suit the application's requirements.
Using a higher grade op amp in the feedback circuit, modifying the compensation
network, and improving the linearity of the triangle waveform should all help
to improve THD and reduce residual noise. In particular, matching the output
filter to the speaker's impedance reduces peaking in the amplifier's
closed-loop response and improves the distortion characteristics of the
amplifier.

Jeffrey D
Sherman is a field applications manager for Harris Semiconductor in Southfield,
MI, and is responsible for the staffing and career development of field
application engineers in the central United States. He holds a bachelor's
degree in electrical engineering from the University of Michigan- Ann Arbor,
MI, and a master's in electrical engineering from Northeastern University in
Boston.

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